REV. A

OP285

–9–

+15V

+

2

3

8

1

4

V

IN

V

OUT

–15V

10pF

+

10F

0.1F

4.99k

2k

0.1F

10F

2.49k

4.99k

+

–

1/2

OP285

Figure 9. Unity-Gain Inverter

In inverting and noninverting applications, the feedback resis-

tance forms a pole with the source resistance and capacitance

(R

S

and C

S

) and the OP285’s input capacitance (C

IN

), as

shown in Figure 10. With R

S

and R

F

in the kilohm range, this

pole can create excess phase shift and even oscillation. A small

capacitor, C

FB

, in parallel with R

FB

eliminates this problem. By

setting R

S

(C

S

+ C

IN

) = R

FB

C

FB

, the effect of the feedback pole

is completely removed.

C

FB

R

FB

C

IN

V

OUT

R

S

C

S

Figure 10. Compensating the Feedback Pole

High-Speed, Low-Noise Differential Line Driver

The circuit of Figure 11 is a unique line driver widely used in

industrial applications. With ±18 V supplies, the line driver can

deliver a differential signal of 30 V p-p into a 2.5 kΩ load. The

high slew rate and wide bandwidth of the OP285 combine to

yield a full power bandwidth of 130 kHz while the low noise

front end produces a referred-to-input noise voltage spectral

density of 10 nV/√Hz. The design is a transformerless, balanced

transmission system where output common-mode rejection of

noise is of paramount importance. Like the transformer-based

design, either output can be shorted to ground for unbalanced

line driver applications without changing the circuit gain of 1.

Other circuit gains can be set according to the equation in the

diagram. This allows the design to be easily set to noninverting,

inverting, or differential operation.

2

3

A2

1

3

2

A1

5

6

7

A3

V

IN

V

O1

V

O2

V

O2

– V

O1

= V

IN

R2

2k

A1 = 1/2OP285

A2, A3 = 1/2 OP285

GAIN = SET R2, R4, R5 = R1 AND R, R7, R8 = R2

1

R1

2k

R3

2k

R9

50

R11

1k

P1

10k

R12

1k

R4

2k

R5

2k

R6

2k

R10

50

R8

2k

R7

2k

Figure 11. High-Speed, Low-Noise Differential Line Driver

Low Phase Error Amplifier

The simple amplifier configuration of Figure 12 uses the OP285

and resistors to reduce phase error substantially over a wide

frequency range when compared to conventional amplifier designs.

This technique relies on the matched frequency characteristics

of the two amplifiers in the OP285. Each amplifier in the circuit

has the same feedback network which produces a circuit gain of

10. Since the two amplifiers are set to the same gain and are

matched due to the monolithic construction of the OP285, they

will exhibit identical frequency response. Recall from feedback

theory that a pole of a feedback network becomes a zero in the

loop gain response. By using this technique, the dominant pole

of the amplifier in the feedback loop compensates for the domi-

nant pole of the main amplifier,

1

2

3

A1

7

A2

5

6

R1

549

R2

4.99k

R3

499

V

IN

V

OUT

R5

549

R4

4.99

A1, A2 = 1/2 OP285

Figure 12. Cancellation of A2’s Dominant Pole by A1

REV. A

OP285

–10–

thereby reducing phase error dramatically. This is shown in

Figure 13 where the 10x composite amplifier’s phase response

exhibits less than 1.5° phase shift through 500 kHz. On the other

hand, the single gain stage amplifier exhibits 25° of phase shift

over the same frequency range. An additional benefit of the low

phase error configuration is constant group delay, by virtue of

constant phase shift at all frequencies below 500 kHz. Although

this technique is valid for minimum circuit gains of 10, actual

closed-loop magnitude response must be optimized for the

amplifier chosen.

–20

–45

10k 100k 10M1M

–25

–30

–35

–40

–15

–10

–5

0

START 10,000.000Hz STOP 10,000,000.000Hz

PHASE – Degrees

SINGLE STAGE

AMPLIFIER RESPONSE

LOW PHASE ERROR

AMPLIFIER RESPONSE

Figure 13. Phase Error Comparison

For a more detailed treatment on the design of low phase error

amplifiers, see Application Note AN-107.

Fast Current Pump

A fast, 30 mA current source, illustrated in Figure 14, takes

advantage of the OP285’s speed and high output current drive.

This is a variation of the Howland current source where a sec-

ond amplifier, A2, is used to increase load current accuracy and

output voltage compliance. With supply voltages of ±15 V, the

output voltage compliance of the current pump is ±8 V. To

keep the output resistance in the MΩ range requires that 0.1%

or better resistors be used in the circuit. The gain of the current

pump can be easily changed according to the equations shown

in the diagram.

1

2

3

A1

5

6

7

V

IN1

V

IN2

A2

A1, A2 = 1/2 OP285

R2

R1

GAIN = , R4 = R2, R3 = R1

R1

2k

R2

2k

R5

50

R3

2k

R4

2k

I

OUT

=

V

IN2

– V

IN1

R5

V

IN

R5

=

I

OUT

= (MAX) = 30mA

Figure 14. A Fast Current Pump

A Low Noise, High Speed Instrumentation Amplifier

A high speed, low noise instrumentation amplifier, constructed

with a single OP285, is illustrated in Figure 15. The circuit exhibits

less than 1.2 µV p-p noise (RTI) in the 0.1 Hz to 10 Hz band

and an input noise voltage spectral density of 9 nV/√Hz (1 kHz)

at a gain of 1000. The gain of the amplifier is easily set by R

G

according to the formula:

V

V

k

R

OUT

IN G

=+

998

2

. Ω

The advantages of a two op amp instrumentation amplifier

based on a dual op amp is that the errors in the individual am-

plifiers tend to cancel one another. For example, the circuit’s

input offset voltage is determined by the input offset voltage

matching of the OP285, which is typically less than 250 µV.

1

2

3

A2

A1

5

6

7

V

IN

A1, A2 = 1/2 OP285

R

Q

9.98k

+2

GAIN =

R

G

()

OPEN

1.24k

102

10

2

10

100

1000

GAIN

R1

4.99k

P1

500

DC CMRR TRIM

AC CMRR TRIM

C1

5pF–40pF

+

–

R

G

R2

4.99

R3

4.99k

R4

4.99k

V

OUT

Figure 15. A High-Speed Instrumentation Amplifier

Common-mode rejection of the circuit is limited by the matching

of resistors R1 to R4. For good common-mode rejection, these

resistors ought to be matched to better than 1%. The circuit was

constructed with 1% resistors and included potentiometer P1

for trimming the CMRR and a capacitor C1 for trimming the

CMRR. With these two trims, the circuit’s common-mode

rejection was better than 95 dB at 60 Hz and better than 65 dB

at 10 kHz. For the best common-mode rejection performance,

use a matched (better than 0.1%) thin-film resistor network for

R1 through R4 and use the variable capacitor to optimize the

circuit’s CMR.

The instrumentation amplifier exhibits very wide small- and

large-signal bandwidths regardless of the gain setting, as shown

in the table. Because of its low noise, wide gain-bandwidth

product, and high slew rate, the OP285 is ideally suited for high

speed signal conditioning applications.

Circuit R

G

Circuit Bandwidth

Gain () V

OUT

= 100 mV p-p V

OUT

= 20 V p-p

2 Open 5 MHz 780 kHz

10 1.24 k 1 MHz 460 kHz

100 102 90 kHz 85 kHz

1000 10 10 kHz 10 kHz

REV. A

OP285

–11–

A 3-Pole, 40 kHz Low-Pass Filter

The closely matched and uniform ac characteristics of the OP285

make it ideal for use in GIC (Generalized Impedance Converter)

and FDNR (Frequency Dependent Negative Resistor) filter appli-

cations. The circuit in Figure 16 illustrates a linear-phase,

3-pole, 40 kHz low-pass filter using an OP285 as an inductance

simulator (gyrator). The circuit uses one OP285 (A2 and A3)

for the FDNR and one OP285 (Al and A4) as an input buffer

and bias current source for A3. Amplifier A4 is configured in a

gain of 2 to set the pass band magnitude response to 0 dB. The

benefits of this filter topology over classical approaches are

that the op amp used in the FDNR is not in the signal path and

that the filter’s performance is relatively insensitive to compo-

nent variations. Also, the configuration is such that large signal

levels can be handled without overloading any of the filter’s

internal nodes. As shown in Figure 17, the OP285’s symmetric

slew rate and low distortion produce a clean, well-behaved

transient response.

10

90

100

0%

SCALE: VERTICAL – 2V/ DIV

HORIZONTAL – 10S/ DIV

V

OUT

10V p-p

10kHz

Figure 17. Low-Pass Filter Transient Response

V

IN

3

2

1

A1

A1, A4 = 1/2 OP285

A2, A3 = 1/2 OP285

1

A2

2

3

R1

95.3k

C1

2200pF

R2

787

C2

2200pF

R3

1.82k

C3

2200pF

R4

1.87k

R5

1.82k

A3

5

7

6

A4

5

7

6

R6

4.12k

R7

100k

R9

1k

V

OUT

R8

1k

C4

2200pF

Figure 16. A 3-Pole, 40 kHz Low-Pass Filter

Driving Capacitive Loads

The OP285 was designed to drive both resistive loads to 600 Ω

and capacitive loads of over 1000 pF and maintain stability. While

there is a degradation in bandwidth when driving capacitive loads,

the designer need not worry about device stability. The graph in

Figure 18 shows the 0 dB bandwidth of the OP285 with capacitive

loads from 10 pF to 1000 pF.

0

0

C

LOAD

– pF

BANDWIDTH – MHz

200 400 600 800 1000

1

2

3

4

5

6

7

8

9

10

Figure 18. Bandwidth vs. C

LOAD

Mfr. #:

Manufacturer:

Analog Devices Inc.

Description:

Precision Amplifiers 9MHz Prec Dual 5mA 250uV

Lifecycle:

New from this manufacturer.

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