 REV. A
OP285
–9–
+15V
+
2
3
8
1
4
V
IN
V
OUT
15V
10pF
+
10F
0.1F
4.99k
2k
0.1F
10F
2.49k
4.99k
+
1/2
OP285
Figure 9. Unity-Gain Inverter
In inverting and noninverting applications, the feedback resis-
tance forms a pole with the source resistance and capacitance
(R
S
and C
S
) and the OP285’s input capacitance (C
IN
), as
shown in Figure 10. With R
S
and R
F
in the kilohm range, this
pole can create excess phase shift and even oscillation. A small
capacitor, C
FB
, in parallel with R
FB
eliminates this problem. By
setting R
S
(C
S
+ C
IN
) = R
FB
C
FB
, the effect of the feedback pole
is completely removed.
C
FB
R
FB
C
IN
V
OUT
R
S
C
S
Figure 10. Compensating the Feedback Pole
High-Speed, Low-Noise Differential Line Driver
The circuit of Figure 11 is a unique line driver widely used in
industrial applications. With ±18 V supplies, the line driver can
deliver a differential signal of 30 V p-p into a 2.5 k load. The
high slew rate and wide bandwidth of the OP285 combine to
yield a full power bandwidth of 130 kHz while the low noise
front end produces a referred-to-input noise voltage spectral
density of 10 nV/Hz. The design is a transformerless, balanced
transmission system where output common-mode rejection of
noise is of paramount importance. Like the transformer-based
design, either output can be shorted to ground for unbalanced
line driver applications without changing the circuit gain of 1.
Other circuit gains can be set according to the equation in the
diagram. This allows the design to be easily set to noninverting,
inverting, or differential operation.
2
3
A2
1
3
2
A1
5
6
7
A3
V
IN
V
O1
V
O2
V
O2
V
O1
= V
IN
R2
2k
A1 = 1/2OP285
A2, A3 = 1/2 OP285
GAIN = SET R2, R4, R5 = R1 AND R, R7, R8 = R2
1
R1
2k
R3
2k
R9
50
R11
1k
P1
10k
R12
1k
R4
2k
R5
2k
R6
2k
R10
50
R8
2k
R7
2k
Figure 11. High-Speed, Low-Noise Differential Line Driver
Low Phase Error Amplifier
The simple amplifier configuration of Figure 12 uses the OP285
and resistors to reduce phase error substantially over a wide
frequency range when compared to conventional amplifier designs.
This technique relies on the matched frequency characteristics
of the two amplifiers in the OP285. Each amplifier in the circuit
has the same feedback network which produces a circuit gain of
10. Since the two amplifiers are set to the same gain and are
matched due to the monolithic construction of the OP285, they
will exhibit identical frequency response. Recall from feedback
theory that a pole of a feedback network becomes a zero in the
loop gain response. By using this technique, the dominant pole
of the amplifier in the feedback loop compensates for the domi-
nant pole of the main amplifier,
1
2
3
A1
7
A2
5
6
R1
549
R2
4.99k
R3
499
V
IN
V
OUT
R5
549
R4
4.99
A1, A2 = 1/2 OP285
Figure 12. Cancellation of A2’s Dominant Pole by A1 REV. A
OP285
–10–
thereby reducing phase error dramatically. This is shown in
Figure 13 where the 10x composite amplifier’s phase response
exhibits less than 1.5° phase shift through 500 kHz. On the other
hand, the single gain stage amplifier exhibits 25° of phase shift
over the same frequency range. An additional benefit of the low
phase error configuration is constant group delay, by virtue of
constant phase shift at all frequencies below 500 kHz. Although
this technique is valid for minimum circuit gains of 10, actual
closed-loop magnitude response must be optimized for the
amplifier chosen.
20
45
10k 100k 10M1M
25
30
35
40
15
10
5
0
START 10,000.000Hz STOP 10,000,000.000Hz
PHASE Degrees
SINGLE STAGE
AMPLIFIER RESPONSE
LOW PHASE ERROR
AMPLIFIER RESPONSE
Figure 13. Phase Error Comparison
For a more detailed treatment on the design of low phase error
amplifiers, see Application Note AN-107.
Fast Current Pump
A fast, 30 mA current source, illustrated in Figure 14, takes
advantage of the OP285’s speed and high output current drive.
This is a variation of the Howland current source where a sec-
ond amplifier, A2, is used to increase load current accuracy and
output voltage compliance. With supply voltages of ±15 V, the
output voltage compliance of the current pump is ±8 V. To
keep the output resistance in the M range requires that 0.1%
or better resistors be used in the circuit. The gain of the current
pump can be easily changed according to the equations shown
in the diagram.
1
2
3
A1
5
6
7
V
IN1
V
IN2
A2
A1, A2 = 1/2 OP285
R2
R1
GAIN = , R4 = R2, R3 = R1
R1
2k
R2
2k
R5
50
R3
2k
R4
2k
I
OUT
=
V
IN2
V
IN1
R5
V
IN
R5
=
I
OUT
= (MAX) = 30mA
Figure 14. A Fast Current Pump
A Low Noise, High Speed Instrumentation Amplifier
A high speed, low noise instrumentation amplifier, constructed
with a single OP285, is illustrated in Figure 15. The circuit exhibits
less than 1.2 µV p-p noise (RTI) in the 0.1 Hz to 10 Hz band
and an input noise voltage spectral density of 9 nV/Hz (1 kHz)
at a gain of 1000. The gain of the amplifier is easily set by R
G
according to the formula:
V
V
k
R
OUT
IN G
=+
998
2
.
The advantages of a two op amp instrumentation amplifier
based on a dual op amp is that the errors in the individual am-
plifiers tend to cancel one another. For example, the circuits
input offset voltage is determined by the input offset voltage
matching of the OP285, which is typically less than 250 µV.
1
2
3
A2
A1
5
6
7
V
IN
A1, A2 = 1/2 OP285
R
Q
9.98k
+2
GAIN =
R
G
()
OPEN
1.24k
102
10
2
10
100
1000
GAIN
R1
4.99k
P1
500
DC CMRR TRIM
AC CMRR TRIM
C1
5pF40pF
+
R
G
R2
4.99
R3
4.99k
R4
4.99k
V
OUT
Figure 15. A High-Speed Instrumentation Amplifier
Common-mode rejection of the circuit is limited by the matching
of resistors R1 to R4. For good common-mode rejection, these
resistors ought to be matched to better than 1%. The circuit was
constructed with 1% resistors and included potentiometer P1
for trimming the CMRR and a capacitor C1 for trimming the
CMRR. With these two trims, the circuits common-mode
rejection was better than 95 dB at 60 Hz and better than 65 dB
at 10 kHz. For the best common-mode rejection performance,
use a matched (better than 0.1%) thin-film resistor network for
R1 through R4 and use the variable capacitor to optimize the
circuits CMR.
The instrumentation amplifier exhibits very wide small- and
large-signal bandwidths regardless of the gain setting, as shown
in the table. Because of its low noise, wide gain-bandwidth
product, and high slew rate, the OP285 is ideally suited for high
speed signal conditioning applications.
Circuit R
G
Circuit Bandwidth
Gain () V
OUT
= 100 mV p-p V
OUT
= 20 V p-p
2 Open 5 MHz 780 kHz
10 1.24 k 1 MHz 460 kHz
100 102 90 kHz 85 kHz
1000 10 10 kHz 10 kHz REV. A
OP285
–11–
A 3-Pole, 40 kHz Low-Pass Filter
The closely matched and uniform ac characteristics of the OP285
make it ideal for use in GIC (Generalized Impedance Converter)
and FDNR (Frequency Dependent Negative Resistor) filter appli-
cations. The circuit in Figure 16 illustrates a linear-phase,
3-pole, 40 kHz low-pass filter using an OP285 as an inductance
simulator (gyrator). The circuit uses one OP285 (A2 and A3)
for the FDNR and one OP285 (Al and A4) as an input buffer
and bias current source for A3. Amplifier A4 is configured in a
gain of 2 to set the pass band magnitude response to 0 dB. The
benefits of this filter topology over classical approaches are
that the op amp used in the FDNR is not in the signal path and
that the filters performance is relatively insensitive to compo-
nent variations. Also, the configuration is such that large signal
internal nodes. As shown in Figure 17, the OP285s symmetric
slew rate and low distortion produce a clean, well-behaved
transient response.
10
90
100
0%
SCALE: VERTICAL 2V/ DIV
HORIZONTAL 10S/ DIV
V
OUT
10V p-p
10kHz
Figure 17. Low-Pass Filter Transient Response
V
IN
3
2
1
A1
A1, A4 = 1/2 OP285
A2, A3 = 1/2 OP285
1
A2
2
3
R1
95.3k
C1
2200pF
R2
787
C2
2200pF
R3
1.82k
C3
2200pF
R4
1.87k
R5
1.82k
A3
5
7
6
A4
5
7
6
R6
4.12k
R7
100k
R9
1k
V
OUT
R8
1k
C4
2200pF
Figure 16. A 3-Pole, 40 kHz Low-Pass Filter
The OP285 was designed to drive both resistive loads to 600
and capacitive loads of over 1000 pF and maintain stability. While
the designer need not worry about device stability. The graph in
Figure 18 shows the 0 dB bandwidth of the OP285 with capacitive
loads from 10 pF to 1000 pF.
0
0
C
pF
BANDWIDTH MHz
200 400 600 800 1000
1
2
3
4
5
6
7
8
9
10
Figure 18. Bandwidth vs. C

#### OP285GSZ

Mfr. #:
Manufacturer:
Analog Devices Inc.
Description:
Precision Amplifiers 9MHz Prec Dual 5mA 250uV
Lifecycle:
New from this manufacturer.
Delivery:
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