AD8628/AD8629/AD8630 Data Sheet
PEAK-TO-PEAK NOISE
Because of the ping-pong action between auto-zeroing and
chopping, the peak-to-peak noise of the AD8628/AD8629/
AD8630 is much lower than the competition. Figure 50 and
Figure 51 show this comparison.
e
n
p-p = 0.5µV
BW = 0.1Hz TO 10Hz
TIME (1s/DIV)
VOLTAGE (0.5µV/DIV)
02735-047
Figure 50. AD8628 Peak-to-Peak Noise
e
n
p-p = 2.3µV
BW = 0.1Hz
T
O 10Hz
TIME (1s/DIV)
VOLT
AGE (0.5µV/DIV)
02735-048
Figure 51. Competitor A Peak-to-Peak Noise
NOISE BEHAVIOR WITH FIRST-ORDER, LOW-PASS
FILTER
The AD8628 was simulated as a low-pass filter (see Figure 53)
and then configured as shown in Figure 52. The behavior of the
AD8628 matches the simulated data. It was verified that noise is
rolled off by first-order filtering. Figure 53 and Figure 54 show
the difference between the simulated and actual transfer functions
of the circuit shown in Figure 52.
470pF
OUT
100kΩ
IN
1kΩ
02735-049
Figure 52. First-Order Low-Pass Filter Test Circuit,
×101 Gain and 3 kHz Corner Frequency
FREQUENC
Y
(kHz)
NOISE (dB)
50
45
40
35
30
25
15
10
5
20
0
0 30
60 100
9080
70
50402010
02735-050
Figure 53. Simulation Transfer Function of the Test Circuit in Figure 52
FREQUENCY (kHz)
NOISE (dB)
50
45
40
35
30
25
15
10
5
20
0
0 30 60
1009080
7050402010
02735-051
Figure 54. Actual Transfer Function of the Test Circuit in Figure 52
The measured noise spectrum of the test circuit charted in
Figure 54 shows that noise between 5 kHz and 45 kHz is
successfully rolled off by the first-order filter.
TOTAL INTEGRATED INPUT-REFERRED NOISE FOR
FIRST-ORDER FILTER
For a first-order filter, the total integrated noise from the
AD8628 is lower than the noise of Competitor A.
3dB FILTER BANDWIDTH (Hz)
RMS NOISE (µV)
10
1
0.1
10 100 10k1k
02735-052
COMPETITOR A
AD8551
AD8628
Figure 55. RMS Noise vs. 3 dB Filter Bandwidth in Hz
Rev. K | Page 16 of 24
Data Sheet AD8628/AD8629/AD8630
INPUT OVERVOLTAGE PROTECTION
Although the AD8628/AD8629/AD8630 are rail-to-rail input
amplifiers, care should be taken to ensure that the potential
difference between the inputs does not exceed the supply voltage.
Under normal negative feedback operating conditions, the
amplifier corrects its output to ensure that the two inputs are at
the same voltage. However, if either input exceeds either supply
rail by more than 0.3 V, large currents begin to flow through the
ESD protection diodes in the amplifier.
These diodes are connected between the inputs and each supply
rail to protect the input transistors against an electrostatic discharge
event, and they are normally reverse-biased. However, if the input
voltage exceeds the supply voltage, these ESD diodes can become
forward-biased. Without current limiting, excessive amounts
of current could flow through these diodes, causing permanent
damage to the device. If inputs are subject to overvoltage,
appropriate series resistors should be inserted to limit the diode
current to less than 5 mA maximum.
OUTPUT PHASE REVERSAL
Output phase reversal occurs in some amplifiers when the input
common-mode voltage range is exceeded. As common-mode
voltage is moved outside the common-mode range, the outputs of
these amplifiers can suddenly jump in the opposite direction to
the supply rail. This is the result of the differential input pair
shutting down, causing a radical shifting of internal voltages
that results in the erratic output behavior.
The AD8628/AD8629/AD8630 amplifiers have been carefully
designed to prevent any output phase reversal, provided that
both inputs are maintained within the supply voltages. If one or
both inputs could exceed either supply voltage, a resistor should
be placed in series with the input to limit the current to less than
5 mA. This ensures that the output does not reverse its phase.
OVERLOAD RECOVERY TIME
Many auto-zero amplifiers are plagued by a long overload recovery
time, often in ms, due to the complicated settling behavior of
the internal nulling loops after saturation of the outputs. The
AD8628/AD8629/AD8630 have been designed so that internal
settling occurs within two clock cycles after output saturation
occurs. This results in a much shorter recovery time, less
than 10 µs, when compared to other auto-zero amplifiers. The
wide bandwidth of the AD8628/AD8629/AD8630 enhances
performance when the parts are used to drive loads that inject
transients into the outputs. This is a common situation when an
amplifier is used to drive the input of switched capacitor ADCs.
TIME (500µs/DIV)
VOLTAGE (V)
V
OUT
0V
0V
V
IN
02735-053
CH1 = 50mV/DIV
CH2 = 1V/DIV
A
V
= –50
Figure 56. Positive Input Overload Recovery for the AD8628
TIME (500µs/DIV)
VOLTAGE (V)
V
OUT
0V
0V
V
IN
02735-054
CH1 = 50mV/DIV
CH2 = 1V/DIV
A
V
= –50
Figure 57. Positive Input Overload Recovery for Competitor A
TIME (500µs/DIV)
VOLTAGE (V)
V
OUT
0V
0V
V
IN
02735-055
CH1 = 50mV/DIV
CH2 = 1V/DIV
A
V
= –50
Figure 58. Positive Input Overload Recovery for Competitor B
Rev. K | Page 17 of 24
AD8628/AD8629/AD8630 Data Sheet
TIME (500µs/DIV)
VOLTAGE (V)
V
OUT
0V
0V
V
IN
02735-056
CH1 = 50mV/DIV
CH2 = 1V/DIV
A
V
= –50
Figure 59. Negative Input Overload Recovery for the AD8628
TIME (500µs/DIV)
VOLTAGE (V)
V
OUT
0V
0V
V
IN
02735-057
CH1 = 50mV/DIV
CH2 = 1V/DIV
A
V
= –50
Figure 60. Negative Input Overload Recovery for Competitor A
TIME (500µs/DIV)
VOLTAGE (V)
V
OUT
0V
0V
V
IN
02735-058
CH1 = 50mV/DIV
CH2 = 1V/DIV
A
V
= –50
Figure 61. Negative Input Overload Recovery for Competitor B
The results shown in Figure 56 to Figure 61 are summarized in
Table 5.
Table 5. Overload Recovery Time
Model
Positive Overload
Recovery (µs)
Negative Overload
Recovery (µs)
AD8628 6 9
Competitor A 650 25,000
Competitor B 40,000 35,000
INFRARED SENSORS
Infrared (IR) sensors, particularly thermopiles, are increasingly
being used in temperature measurement for applications as wide
ranging as automotive climate control, human ear thermometers,
home insulation analysis, and automotive repair diagnostics.
The relatively small output signal of the sensor demands high
gain with very low offset voltage and drift to avoid dc errors.
If interstage ac coupling is used, as in Figure 62, low offset and
drift prevent the output of the input amplifier from drifting close to
saturation. The low input bias currents generate minimal errors
from the output impedance of the sensor. As with pressure sensors,
the very low amplifier drift with time and temperature eliminate
additional errors once the temperature measurement is calibrated.
The low 1/f noise improves SNR for dc measurements taken
over periods often exceeding one-fifth of a second.
Figure 62 shows a circuit that can amplify ac signals from 100 µV to
300 µV up to the 1 V to 3 V levels, with a gain of 10,000 for
accurate analog-to-digital conversion.
5V
100kΩ10kΩ
5V
100µV TO 300µV
100Ω
TO BIAS
VOLTAGE
10kΩ
f
C
1.6Hz
IR
DETECTOR
100kΩ
10µF
1/2 AD8629
1/2 AD8629
02735-059
Figure 62. AD8629 Used as Preamplifier for Thermopile
Rev. K | Page 18 of 24

AD8630WARZ-RL

Mfr. #:
Manufacturer:
Analog Devices Inc.
Description:
Precision Amplifiers Zero Drift RRIO Quad SGL-Supply
Lifecycle:
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