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LT3474EFE#PBF

P1-P3P4-P6P7-P9P10-P12P13-P15P16-P18P19-P20
LT3474/LT3474-1
10
3474fd
APPLICATIONS INFORMATION
Setting the Switching Frequency
The LT3474 uses a constant frequency architecture that
can be programmed over a 200kHz to 2MHz range with a
single external timing resistor from the R
T
pin to ground.
The current that fl ows into the timing resistor is used
to charge an internal oscillator capacitor. A graph for
selecting the value of R
T
for a given operating frequency
is shown in the Typical Performance Characteristics
section. Table 1 shows suggested R
T
selections for a
variety of switching frequencies.
Table 1. Switching Frequencies
SWITCHING FREQUENCY (MHz) R
T
(kΩ)
210
1.5 18.7
1 33.2
0.7 52.3
0.5 80.6
0.3 147
0.2 232
Operating Frequency Selection
The choice of operating frequency is determined by sev-
eral factors. There is a tradeoff between effi ciency and
component size. Higher switching frequency allows the
use of smaller inductors at the cost of increased switching
losses and decreased effi ciency.
Another consideration is the maximum duty cycle. In
certain applications, the converter needs to operate at a
high duty cycle in order to work at the lowest input voltage
possible. The LT3474 has a fi xed oscillator off-time and
a variable on-time. As a result, the maximum duty cycle
increases as the switching frequency is decreased.
Input Voltage Range
The minimum operating voltage is determined either by the
LT3474’s undervoltage lockout of 4V, or by its maximum
duty cycle. The duty cycle is the fraction of time that the
internal switch is on and is determined by the input and
output voltages:
DC
VV
VV V
OUT F
IN SW F
=
+
()
+
()
–
where V
F
is the forward voltage drop of the catch diode
(~0.4V) and V
SW
is the voltage drop of the internal switch
(~0.4V at maximum load). This leads to a minimum input
voltage of:
V
VV
DC
VV
IN MIN
OUT F
MAX
FSW
()
=
+
+–
with DC
MAX
= 1–t
OFF(MIN)
• f
where t
0FF(MIN)
is equal to 200ns and f is the switching
frequency.
Example: f = 500kHz, V
OUT
= 4V
DC ns kHz
V
VV
MAX
IN MIN
= − =
=
+
()
1 200 500 0 90
404
0
•.
.
.
99
04 04 49–. . .VVV+=
The maximum operating voltage is determined by the
absolute maximum ratings of the V
IN
and BOOST pins,
and by the minimum duty cycle.
V
VV
DC
VV
IN MAX
OUT F
MIN
FSW
()
=
+
+–
with DC
MIN
= t
ON(MIN)
• f
where t
ON(MIN)
is equal to 160ns and f is the switching
frequency.
Example: f = 500kHz, V
OUT
= 2.5V
DC ns kHz
V
VV
MIN
IN MAX
==
=
+
()
160 500 0 08
25 04
0
•.
..
.
008
04 04 36–. .VVV+=
The minimum duty cycle depends on the switching fre-
quency. Running at a lower switching frequency might
allow a higher maximum operating voltage. Note that this
is a restriction on the operating input voltage; the circuit
will tolerate transient inputs up to the Absolute Maximum
Rating.
LT3474/LT3474-1
11
3474fd
APPLICATIONS INFORMATION
The optimum inductor for a given application may differ
from the one indicated by this simple design guide. A
larger value inductor provides a higher maximum load
current, and reduces the output voltage ripple. If your
load is lower than the maximum load current, then you
can relax the value of the inductor and operate with higher
ripple current. This allows you to use a physically smaller
inductor, or one with a lower DCR resulting in higher
effi ciency. Be aware that if the inductance differs from
the simple rule above, then the maximum load current
will depend on input voltage. In addition, low inductance
may result in discontinuous mode operation, which further
reduces maximum load current. For details of maximum
output current and discontinuous mode operation, see
Linear Technology’s Application Note 44. Finally, for duty
cycles greater than 50% (V
OUT
/V
IN
> 0.5), a minimum
inductance is required to avoid sub-harmonic oscillations.
See Application Note 19.
The current in the inductor is a triangle wave with an average
value equal to the load current. The peak switch current
is equal to the output current plus half the peak-to-peak
inductor ripple current. The LT3474 limits its switch cur-
rent in order to protect itself and the system from overload
faults. Therefore, the maximum output current that the
LT3474 will deliver depends on the switch current limit,
the inductor value, and the input and output voltages.
When the switch is off, the potential across the inductor
is the output voltage plus the catch diode drop. This gives
the peak-to-peak ripple current in the inductor
ΔI
L
=
1–DC
()
V
OUT
+ V
F
()
L•f
()
where f is the switching frequency of the LT3474 and L
is the value of the inductor. The peak inductor and switch
current is
I
SW PK
()
=I
LPK
()
=I
OUT
+
ΔI
L
2
Inductor Selection and Maximum Output Current
A good first choice for the inductor value is
LV V
kHz
f
OUT F
=+•()
900
where V
F
is the voltage drop of the catch diode (~0.4V), f
is the switching frequency and L is in μH. With this value
the maximum load current will be 1.1A, independent of
input voltage. The inductor’s RMS current rating must be
greater than the maximum load current and its saturation
current should be at least 30% higher. For highest effi ciency,
the series resistance (DCR) should be less than 0.2Ω.
Table 2 lists several vendors and types that are suitable.
For robust operation at full load and high input voltages
(V
IN
> 30V), use an inductor with a saturation current
higher than 2.5A.
Table 2. Inductors
PART NUMBER
VALUE
(μH)
I
RMS
(A)
DCR
(Ω)
HEIGHT
(mm)
Sumida
CR43-3R3 3.3
1.44 0.086 3.5
CR43-4R7 4.7 1.15 0.109 3.5
CDRH4D16-3R3 3.3 1.1 0.063 1.8
CDRH4D28-3R3 3.3 1.57 0.049 3
CDRH4D28-4R7 4.7 1.32 0.072 3
CDRH5D28-100 10 1.3 0.048 3
CDRH5D28-150 15 1.1 0.076 3
CDRH73-100 10 1.68 0.072 3.4
CDRH73-150 15 1.33 0.13 3.4
Coilcraft
DO1606T-332
3.3 1.3 0.1 2
DO1606T-472 4.7 1.1 0.12 2
DO1608C-332 3.3 2 0.08 2.9
DO1608C-472 4.7 1.5 0.09 2.9
MOS6020-332 3.3 1.8 0.046 2
MOS6020-472 10 1.5 0.05 2
LT3474/LT3474-1
12
3474fd
APPLICATIONS INFORMATION
To maintain output regulation, this peak current must be
less than the LT3474’s switch current limit I
LIM
. For SW1,
I
LIM
is at least 1.6A (1.5A at 125°C) at low duty cycles and
decreases linearly to 1.15A (1.08A at 125°C) at DC = 0.8.
The maximum output current is a function of the chosen
inductor value:
I
OUT MAX
()
= I
LIM
–
ΔI
L
2
=1.6A • 1– 0.35•DC
()
–
ΔI
L
2
Choosing an inductor value so that the ripple current is
small will allow a maximum output current near the switch
current limit.
One approach to choosing the inductor is to start with the
simple rule given above, look at the available inductors,
and choose one to meet cost or space goals. Then use
these equations to check that the LT3474 will be able to
deliver the required output current. Note again that these
equations assume that the inductor current is continuous.
Discontinuous operation occurs when I
OUT
is less than
ΔI
L
/2.
Input Capacitor Selection
Bypass the input of the LT3474 circuit with a 2.2μF or
higher ceramic capacitor of X7R or X5R type. A lower
value or a less expensive Y5V type will work if there is
additional bypassing provided by bulk electrolytic capaci-
tors or if the input source impedance is low. The following
paragraphs describe the input capacitor considerations in
more detail.
Step-down regulators draw current from the input sup-
ply in pulses with very fast rise and fall times. The input
capacitor is required to reduce the resulting voltage ripple
at the LT3474 input and to force this switching current into
a tight local loop, minnimizing EMI. The input capacitor
must have low impedance at the switching frequency to
do this effectively, and it must have an adequate ripple
current rating. The RMS input is:
CI
VVV
V
I
INRMS OUT
OUT IN OUT
IN
OUT
=
()
<•
–
2
and is largest when V
IN
= 2V
OUT
(50% duty cycle). Con-
sidering that the maximum load current is 1A, RMS ripple
current will always be less than 0.5A
The high switching frequency of the LT3474 reduces the
energy storage requirements of the input capacitor, so that
the capacitance required is less than 10μF. The combination
of small size and low impedance (low equivalent series
resistance or ESR) of ceramic capacitors makes them the
preferred choice. The low ESR results in very low voltage
ripple. Ceramic capacitors can handle larger magnitudes
of ripple current than other capacitor types of the same
value. Use X5R and X7R types.
An alternative to a high value ceramic capacitor is a lower
value ceramic along with a larger electrolytic capaci-
tor. The electrolytic capacitor likely needs to be greater
than 10μF in order to meet the ESR and ripple current
requirements. The input capacitor is likely to see high
surge currents when the input source is applied. Tanta-
lum capacitors can fail due to an over-surge of current.
Only use tantalum capacitors with the appropriate surge
current rating. The manufacturer may also recommend
operation below the rated voltage of the capacitor.
P1-P3P4-P6P7-P9P10-P12P13-P15P16-P18P19-P20

LT3474EFE#PBF

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Manufacturer:
Analog Devices / Linear Technology
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LED Lighting Drivers 1A Step Down LED Driver in TSSOP-16
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