Data Sheet AD8209A
Rev. A | Page 13 of 15
GAIN TRIM
Figure 31 shows a method for incremental gain trimming by
using a trim potentiometer and an external resistor, R
EXT
.
The following approximation is useful for small gain ranges:
ΔG ≈ (10 MΩ ÷ R
EXT
)%
For example, using this equation, the adjustment range is ±2%
for R
EXT
= 5 MΩ and ±10% for R
EXT
= 1 MΩ.
Figure 31. Incremental Gain Trimming
Internal Signal Overload Considerations
When configuring the gain for values other than 14, the maximum
input voltage with respect to the supply voltage and ground must
be considered because either the preamplifier or the output buffer
reaches its full-scale output (V
S
− 0.1 V) with large differential input
voltages. The input of the AD8209A is limited to (V
S
− 0.1) ÷ 7 for
overall gains of ≤7 because the preamplifier, with its fixed gain of
7 V/V, reaches its full-scale output before the output buffer. For
gains greater than 7, the swing at the buffer output reaches its full
scale first and then limits the AD8209A input to (V
S
− 0.1) ÷ G,
where G is the overall gain.
LOW-PASS FILTERING
In many transducer applications, it is necessary to filter the signal
to remove spurious high frequency components, including noise,
or to extract the mean value of a fluctuating signal with a peak
to average ratio (PAR) greater than unity. For example, a full wave
rectified sinusoid has a PAR of 1.57, a raised cosine has a PAR
of 2, and a half wave sinusoid has a PAR of 3.14. Signals with
large spikes can have PARs of 10 or more.
When implementing a filter, consider the PAR so that the output of
the AD8209A preamplifier (A1) does not clip before A2; otherwise,
the nonlinearity is averaged and appears as an error at the output.
To avoid this error, both amplifiers clip at the same time. This
condition is achieved when the PAR is no greater than the gain
of the second amplifier (2 for the default configuration). For
example, if a PAR of 5 is expected, increase the gain of A2 to 5.
Low-pass filters can be implemented in several ways by using
the features provided by the AD8209A. In the simplest case, a
single-pole filter (20 dB/decade) is formed when the output
of A1 is connected to the input of A2 via the internal 100 kΩ
resistor by tying Pin 3 to Pin 4 and adding a capacitor from this
node to ground, as shown in Figure 32. If a resistor is added
across the capacitor to lower the gain, the corner frequency
increases; therefore, calculate the gain using the parallel sum
of the resistor and 100 kΩ.
Figure 32. Single-Pole, Low-Pass Filter Using the Internal 100 kΩ Resistor
If the gain is raised using a resistor, as shown in Figure 30, the
corner frequency is lowered by the same factor as the gain is raised.
Therefore, using a resistor of 200 kΩ (for which the gain is
doubled), results in a corner frequency scaled to 0.796 Hz/µF
(0.039 µF for a 20 Hz corner frequency).
Figure 33. Two-Pole, Low-Pass Filter
A two-pole filter with a roll-off of 40 dB/decade can be
implemented using the connections shown in Figure 33. This
configuration is a Sallen-Key form based on a ×2 amplifier. A
two-pole filter with a corner frequency of f
2
and a single-pole filter
with a corner frequency of f
1
have the same attenuation, that is,
40 log (f
2
/f
1
), as shown in Figure 34. Using the standard resistor
value shown in Figure 33 and capacitors of equal values, the corner
frequency is conveniently scaled to 1 Hz/µF (0.05 µF for a 20 Hz
corner frequency). A maximal flat response occurs when the
resistor is lowered to 196 kΩ, scaling the corner frequency to
GND
DNC
–IN
+IN
A1
V
S
A2
R
EXT
OUT
AD8209A
5V
V
DIFF
V
CM
DNC = DO NOT CONNECT
OUTPUT
GAIN TRIM
20kΩ MIN
+
–
+
–
14511-033
GND
DNC
–IN
+IN
A1
V
S
A2
OUT
AD8209
A
5V
V
DIFF
V
CM
C
F
DNC = DO NOT CONNECT
OUTPUT
f
C
=
1
2πC10
5
C IN FARADS
+
–
+
–
145
11-034
GND
DNC
–IN
+IN
A1
V
S
A2
OUT
AD8209A
5V
V
DIFF
V
CM
C
C
DNC = DO NOT CONNECT
OUTPUT
f
C
(Hz) = 1/C(µF)
255kΩ
+
–
+
–
14511-035
