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MAX4223EUT+T

P1-P3P4-P6P7-P9P10-P12P13-P15P16-P18P19-P20
MAX4223–MAX4228
1GHz, Low-Power, SOT23,
Current-Feedback Amplifiers with Shutdown
10 ______________________________________________________________________________________
_______________Detailed Description
The MAX4223–MAX4228 are ultra-high-speed, low-
power, current-feedback amplifiers featuring -3dB
bandwidths up to 1GHz, 0.1dB gain flatness up to
300MHz, and very low differential gain and phase
errors of 0.01% and 0.02°, respectively. These devices
operate on dual ±5V or ±3V power supplies and
require only 6mA of supply current per amplifier. The
MAX4223/MAX4225/MAX4226 are optimized for
closed-loop gains of +1 (0dB) or more and have -3dB
bandwidths of 1GHz. The MAX4224/MAX4227/
MAX4228 are optimized for closed-loop gains of +2
(6dB) or more, and have -3dB bandwidths of 600MHz
(1.2GHz gain-bandwidth product).
The current-mode feedback topology of these ampli-
fiers allows them to achieve slew rates of up to
1700V/µs with corresponding large signal bandwidths
up to 330MHz. Each device in this family has an output
that is capable of driving a minimum of 60mA of output
current to ±2.5V.
Theory of Operation
Since the MAX4223–MAX4228 are current-feedback
amplifiers, their open-loop transfer function is
expressed as a transimpedance:
The frequency behavior of this open-loop transimped-
ance is similar to the open-loop gain of a voltage-feed-
back amplifier. That is, it has a large DC value and
decreases at approximately 6dB per octave.
Analyzing the current-feedback amplifier in a gain con-
figuration (Figure 1) yields the following transfer func-
tion:
At low gains, (G x R
IN-
) << R
F
. Therefore, unlike tradi-
tional voltage-feedback amplifiers, the closed-loop
bandwidth is essentially independent of the closed-
loop gain. Note also that at low frequencies, T
Z
>> [(G
x R
IN-
) + R
F
], so that:
Low-Power Shutdown Mode
The MAX4223/MAX4224/MAX4226/MAX4228 have a
shutdown mode that is activated by driving the SHDN
input low. When powered from ±5V supplies, the SHDN
input is compatible with TTL logic. Placing the amplifier
in shutdown mode reduces quiescent supply current to
350µA typical, and puts the amplifier output into a high-
impedance state (100kΩ typical). This feature allows
these devices to be used as multiplexers in wideband
systems. To implement the mux function, the outputs of
multiple amplifiers can be tied together, and only the
amplifier with the selected input will be enabled. All of
the other amplifiers will be placed in the low-power
shutdown mode, with their high output impedance pre-
senting very little load to the active amplifier output. For
gains of +2 or greater, the feedback network imped-
ance of all the amplifiers used in a mux application
must be considered when calculating the total load on
the active amplifier output.
__________Applications Information
Layout and Power-Supply Bypassing
The MAX4223–MAX4228 have an extremely high band-
width, and consequently require careful board layout,
including the possible use of constant-impedance
microstrip or stripline techniques.
V
V
G
R
R
OUT
IN
F
G
= = + 1
.
V
V
G x
T S
T S G x R R
where G A
R
R
OUT
IN
Z
Z IN F
V
F
G
=
( )
( )
+ +
= = +
−
1
∆
∆
V
I
or T
OUT
IN
Z
−
MAX4223
MAX4224
MAX4225
MAX4226
MAX4227
MAX4228
R
G
IN-
T
Z
R
IN-
OUT
+1
IN+
V
IN
R
F
+1
Figure 1. Current-Feedback Amplifier
MAX4223–MAX4228
1GHz, Low-Power, SOT23,
Current-Feedback Amplifiers with Shutdown
______________________________________________________________________________________ 11
To realize the full AC performance of these high-speed
amplifiers, pay careful attention to power-supply
bypassing and board layout. The PC board should
have at least two layers: a signal and power layer on
one side and a large, low-impedance ground plane on
the other. The ground plane should be as free of voids
as possible, with one exception: the inverting input pin
(IN-) should have as low a capacitance to ground as
possible. This means that there should be no ground
plane under IN- or under the components (R
F
and R
G
)
connected to it. With multilayer boards, locate the
ground plane on a layer that incorporates no signal or
power traces.
Whether or not a constant-impedance board is used, it
is best to observe the following guidelines when
designing the board:
1) Do not use wire-wrapped boards (they are too
inductive) or breadboards (they are too capacitive).
2) Do not use IC sockets. IC sockets increase reac-
tance.
3) Keep signal lines as short and straight as possible.
Do not make 90° turns; round all corners.
4) Observe high-frequency bypassing techniques to
maintain the amplifier’s accuracy and stability.
5) In general, surface-mount components have shorter
bodies and lower parasitic reactance, giving better
high-frequency performance than through-hole com-
ponents.
The bypass capacitors should include a 10nF ceramic,
surface-mount capacitor between each supply pin and
the ground plane, located as close to the package as
possible. Optionally, place a 10µF tantalum capacitor
at the power-supply pins’ point of entry to the PC board
to ensure the integrity of incoming supplies. The power-
supply trace should lead directly from the tantalum
capacitor to the V
CC
and V
EE
pins. To minimize para-
sitic inductance, keep PC traces short and use surface-
mount components. The N.C. pins should be
connected to a common ground plane on the PC board
to minimize parasitic coupling.
If input termination resistors and output back-termina-
tion resistors are used, they should be surface-mount
types, and should be placed as close to the IC pins as
possible. Tie all N.C. pins to the ground plane to mini-
mize parasitic coupling.
Choosing Feedback and Gain Resistors
As with all current-feedback amplifiers, the frequency
response of these devices depends critically on the
value of the feedback resistor R
F
. R
F
combines with an
internal compensation capacitor to form the dominant
pole in the feedback loop. Reducing R
F
’s value
increases the pole frequency and the -3dB bandwidth,
but also increases peaking due to interaction with other
nondominant poles. Increasing R
F
’s value reduces
peaking and bandwidth.
Table 1 shows optimal values for the feedback resistor
(R
F
) and gain-setting resistor (R
G
) for the MAX4223–
MAX4228. Note that the MAX4224/MAX4227/MAX4228
offer superior AC performance for all gains except unity
gain (0dB). These values provide optimal AC response
using surface-mount resistors and good layout tech-
niques. Maxim’s high-speed amplifier evaluation kits
provide practical examples of such layout techniques.
Stray capacitance at IN- causes feedback resistor
decoupling and produces peaking in the frequency-
response curve. Keep the capacitance at IN- as low as
possible by using surface-mount resistors and by
avoiding the use of a ground plane beneath or beside
these resistors and the IN- pin. Some capacitance is
unavoidable; if necessary, its effects can be counter-
acted by adjusting R
F
. Use 1% resistors to maintain
consistency over a wide range of production lots.
Table 1. Optimal Feedback Resistor
Networks
MAX4223/MAX4225/MAX4226
2 6 200 200 380 115
GAIN
(dB)
R
G
(Ω)
R
F
(Ω)
0.1dB
BW
(MHz)
GAIN
(V/V)
-3dB
BW
(MHz)
5 14 100 25 235 65
2 6 470 470 600 200
5 14 240 62 400 90
10 20 130 15 195 35
MAX4224/MAX4227/MAX4228
*
For the MAX4223EUT, this optimal value is 470
Ω
.
1 0 560* Open 1000 300
MAX4223–MAX4228
1GHz, Low-Power, SOT23,
Current-Feedback Amplifiers with Shutdown
12 ______________________________________________________________________________________
DC and Noise Errors
The MAX4223–MAX4228 output offset voltage, V
OUT
(Figure 2), can be calculated with the following equation:
where:
V
OS
= input offset voltage (in volts)
1 + R
F
/ R
G
= amplifier closed-loop gain (dimensionless)
I
B+
= input bias current (in amps)
I
B-
= inverting input bias current (in amps)
R
G
= gain-setting resistor (in Ω)
R
F
= feedback resistor (in Ω)
R
S
= source resistor (in Ω)
The following equation represents output noise density:
where:
i
n
= input noise current density (in pA/√Hz)
e
n
= input noise voltage density (in nV/√Hz)
The MAX4223–MAX4228 have a very low, 2nV/√Hz
noise voltage. The current noise at the noninverting
input (i
n+
) is 3pA/√Hz, and the current noise at the
inverting input (i
n-
) is 20pA/√Hz.
An example of DC-error calculations, using the
MAX4224 typical data and the typical operating circuit
with R
F
= R
G
= 470Ω (R
F
|| R
G
= 235Ω) and R
S
= 50Ω,
gives:
V
OUT
= [5 x 10
-4
x (1 + 1)] + [2 x 10
-6
x 50 x (1 + 1)] +
[4 x 10
-6
x 470]
V
OUT
= 3.1mV
Calculating total output noise in a similar manner yields
the following:
With a 600MHz system bandwidth, this calculates to
250µV
RMS
(approximately 1.5mVp-p, using the six-
sigma calculation).
Communication Systems
Nonlinearities of components used in a communication
system produce distortion of the desired output signal.
Intermodulation distortion (IMD) is the distortion that
results from the mixing of two input signals of different
frequencies in a nonlinear system. In addition to the
input signal frequencies, the resulting output signal
contains new frequency components that represent the
sum and difference products of the two input frequen-
cies. If the two input signals are relatively close in fre-
quency, the third-order sum and difference products
will fall close to the frequency of the desired output and
will therefore be very difficult to filter. The third-order
intercept (IP3) is defined as the power level at which
the amplitude of the largest third-order product is equal
to the power level of the desired output signal. Higher
third-order intercept points correspond to better lineari-
ty of the amplifier. The MAX4223–MAX4228 have a typi-
cal IP3 value of 42dBm, making them excellent choices
for use in communications systems.
ADC Input Buffers
Input buffer amplifiers can be a source of significant
errors in high-speed ADC applications. The input buffer
is usually required to rapidly charge and discharge the
ADC’s input, which is often capacitive (see the section
Driving Capacitive Loads
). In addition, a high-speed
ADC’s input impedance often changes very rapidly
during the conversion cycle, requiring an amplifier with
e x
x x
x x x
e nV Hz
n OUT
n OUT
( )
−
− −
( )
= +
( )
+
+
=
. /
1 1
3 10 50
20 10 235 2 10
10 2
12
12
2
9
2
2
e
R
R
x
i x R i x R R e
n OUT
F
G
n S n F G n
( )
+ −
= +
( )
+
( )
[ ]
+
( )
||
1
2
2
2
V V x 1 R /R I x R
x 1
R
R
I x R
OUT OS F G B S
F
G
B F
= +
( )
+
+
+
+
−
MAX4223
MAX4224
MAX4225
MAX4226
MAX4227
MAX4228
R
G
IN-
I
B-
I
B+
IN+
V
OUT
OUT
R
S
R
F
Figure 2. Output Offset Voltage
P1-P3P4-P6P7-P9P10-P12P13-P15P16-P18P19-P20

MAX4223EUT+T

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Manufacturer:
Maxim Integrated
Description:
Operational Amplifiers - Op Amps 1GHz Current Feedback Amp
Lifecycle:
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