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LT3509EMSE#PBF

P1-P3P4-P6P7-P9P10-P12P13-P15P16-P18P19-P21P22-P24
LT3509
13
3509fd
For more information www.linear.com/LT3509
applicaTions inForMaTion
Figure 7. BD Tied to Regulated Output
L1
D1
C
OUT
V
OUT
C
BOOST
V
BOOST
–
V
SW
≅ V
OUT
MAX
V
BOOST
≅ V
IN
+ V
OUT
V
OUT
≥ 3V
V
IN
3509 F07
LT3509
GND
V
IN
BD
BOOST
SW
DA
C
IN
Boost Pin Considerations
Figure 7 through Figure 9 show several ways to arrange
the boost circuit. The BOOST pin must be more than 2V
above the SW pin for full efficiency. For outputs of 3.3V
and higher, the standard circuit Figure 7 is best. For lower
output voltages, the boost diode can be tied to the input
Figure 8. The circuit in Figure 7 is more efficient because
the boost pin current comes from a lower voltage source.
Finally, as shown in Figure 9, the BD pin can be tied to
another source that is at least 3V. For example, if you are
generating 3.3V and 1.8V, and the 3.3V is on whenever
the 1.8V is on, the 1.8V boost diode can be connected to
the 3.3V output.
In any case, be sure that the maximum voltage at the
BOOST pin is less than 60V and the voltage difference
between the BOOST and SW pins is less than 30V.
Inductor Selection and Maximum Output Current
A good first choice for the inductor value is:
LV V
MHz
f
OUT F
SW
=+
()
•
.21
where V
F
is the voltage drop of the catch diode (~0.5V)
and L is in µH.
The inductor’s RMS current rating must be greater than the
maximum load current and its saturation current should
be at least 30% higher. For highest efficiency, the series
resistance (DCR) should be less than 0.15Ω. Table 2 lists
several vendors and types that are suitable.
The current in the inductor is a triangle wave with an average
value equal to the load current. The peak switch current
is equal to the output current plus half the peak-to-peak
inductor ripple current. The LT3509 limits its switch current
in order to protect itself and the system from overcurrent
faults. Therefore, the maximum output current that the
LT3509 will deliver depends on the switch current limit,
the inductor value and the input and output voltages.
L1
D1
C
OUT
V
OUT
C
BOOST
V
BOOST
–
V
SW
≅ V
IN
MAX
V
BOOST
≅ 2V
IN
C
IN
V
IN
3509 F08
LT3509
GND
V
IN
BD
BOOST
SW
DA
Figure 8. Supplied from V
IN
L1
D1
C
OUT
V
OUT
C
BOOST
V
BOOST
–
V
SW
≅ V
BD
MAX
V
BOOST
≅ V
IN
+ V
BD
V
BD
≥ 3V
C
IN
V
IN
V
BD
3509 F09
LT3509
GND
V
IN
BD
BOOST
SW
DA
Figure 9. Separate Boost Supply
LT3509
14
3509fd
For more information www.linear.com/LT3509
When the switch is off, the potential across the inductor is
the output voltage plus the catch diode forward voltage. This
gives the peak-to-peak ripple current in the inductor:
∆I
L
=(1– DC)
V
OUT
+
V
F
L • f
SW
where:
DC = Duty Cycle
f
SW
= Switching Frequency
L = Inductor Value
V
F
= Diode Forward Voltage
The peak inductor and switch current is:
I
SWPK
=I
LPK
=I
OUT
+
∆
I
L
2
To maintain output regulation, this peak current must be
less than the LT3509’s switch current limit I
LIM
. This is
dependent on duty cycle due to the slope compensation.
For I
LIM
is at least 1.4A at low duty cycles and decreases
linearly to 1.0A at DC = 0.8.
The theoretical minimum inductance can now be calcu
-
lated as:
L
DC
f
VV
II
MIN
MIN
OUT F
LIM OUT
=
+
1–
•
–
where DC
MIN
is the minimum duty cycle called for by the
application i.e.:
DC
VV
VV
V
MIN
OUT MA
XF
IN MINS
WF
=
+
+
()
()
–
There is a limit to the actual minimum duty cycle imposed
by the minimum on-time of the switch. For a robust design
it is important that inductor that will not saturate when
the switch is at its minimum on-time, the input voltage
is at maximum and the output is short circuited. In this
case the full input voltage, less the drop in the switch, will
appear across the inductor. This doesn’t require an actual
short, just starting into a capacitive load will provide the
same conditions. The Diode current sensing scheme will
ensure that the switch will not turn-on if the inductor
current is above the DA current limit threshold, which has
a maximum of 1.1A. The peak current under short-circuit
conditions can then be calculated from:
I
Vt
L
A
PEAK
IN ON MIN
=+
•
.
()
11
The inductor should have a saturation current greater than
this value. For safe operation with high input voltages this
can often mean using a physically larger inductor as higher
value inductors often have lower saturation currents for
a given core size. As a general rule the saturation current
should be at least 1.8A to be short-circuit proof. However, it’s
generally better to use an inductor larger than the minimum
value. For robust operation at input voltages greater than
30V, use an inductor with a value of 4.2µH or greater, and
a saturation current rating of 1.8A or higher. The minimum
inductor has large ripple currents which increase core
losses and require large output capacitors to keep output
voltage ripple low. Select an inductor greater than L
MIN
that keeps the ripple current below 30% of I
LIM
.
applicaTions inForMaTion
LT3509
15
3509fd
For more information www.linear.com/LT3509
applicaTions inForMaTion
Table 2. Recommended Inductors
MANUFACTURER/
PART NUMBER
VALUE
(µH)
I
SAT
(A)
DCR
(W)
HEIGHT
(mm)
Coilcraft
LPS4018-222ML 2.2 2.8 0.07 1.7
LPS5030-332ML 3.3 2.5 0.066 2.9
LPS5030-472ML 4.7 2.5 0.083 2.9
LPS6225-682ML 6.8 2.7 0.095 2.4
LPS6225-103ML 10 2.1 0.105 2.4
Sumida
CDRH4D22/HP-2R2N 2.2 3.2 0.0035 2.4
CDRH4D22/HP-3R5N 3.5 2.5 0.052 2.4
CDRH4D22/HP-4R7N 4.7 2.2 0.066 2.4
CDRH5D28/HP-6R8N 6.8 3.1 0.049 3.0
CDRH5D28/HP-8R2N 8.2 2.7 0.071 3.0
CDRH5D28R/HP-100N 10 2.45 0.074 3.0
Cooper
SD52-2R2-R 2.2 2.30 0.0385 2.0
SD52-3R5-R 3.5 1.82 0.0503 2.0
SD52-4R7-R 4.7 1.64 0.0568 2.0
SD6030-5R8-R 5.8 1.8 0.045 3.0
SD7030-8R0-R 8.0 1.85 0.058 3.0
SD7030-100-R 10.0 1.7 0.065 3.0
Toko
A997AS-2R2N 2.2
1.6 0.06 1.8
A997AS-3R3N 3.3 1.2 0.07 1.8
A997AS-4R7M 4.7 1.07 0.1 1.8
Würth
7447745022 2.2 3.5 0.036 2.0
7447745033 3.3 3.0 0.045 2.0
7447745047 4.7 2.4 0.057 2.0
7447745076 7.6 1.8 0.095 2.0
7447445100 10 1.6 0.12 2.0
The prior analysis is valid for continuous mode operation
(I
OUT
> ∆I
LIM
/ 2). For details of maximum output current
in discontinuous mode operation, see Linear Technology’s
Application Note 44. Finally, for duty cycles greater than
50% (V
OUT
/V
IN
> 0.5), a minimum inductance is required
to avoid subharmonic oscillations. This minimum induc-
tance is
LVV
f
MIN OUT F
SW
=+
()
•
.14
where f
SW
is in MHz and L
MIN
is in µH.
If using external synchronization, calculate L
MIN
using the
R
T
frequency and not the SYNC frequency.
Frequency Compensation
The LT3509 uses current mode control to regulate the
output, which simplifies loop compensation and allows
the necessary filter components to be integrated. The fixed
internal compensation network has been chosen to give
stable operation over a wide range of operating conditions
but assumes a minimum load capacitance. The LT3509
does not depend on the ESR of the output capacitor for
stability so the designer is free to use ceramic capacitors
to achieve low output ripple and small PCB footprint.
Figure 10 shows an equivalent circuit for the LT3509 control
loop. The error amp is a transconductance amplifier with
finite output impedance. The power section, consisting of
the modulator, power switch and inductor is modeled as a
transconductance amplifier generating an output current
proportional to the voltage at the COMP-NODE. The gain
of the power stage (gmp) is 1.1S. Note that the output
capacitor integrates this current and that the internal
capacitor integrates the error amplifier output current,
resulting in two poles in the loop. In most cases, a zero is
required and comes either from the output capacitor ESR
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LT3509EMSE#PBF

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Buy LT3509EMSE#PBF
Manufacturer:
Analog Devices / Linear Technology
Description:
Switching Voltage Regulators Dual Integrated 700mA Wide Input Rane Step-Down Regulator
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