LTC3114HDHC-1#TRPBF Datasheet LTC3114EDHC-1#TRPBF, LTC3114MPDHC-1#PBF, LTC3114MPFE-1#PBF | P13-P15

LTC3114-1

31141fb

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operaTion

sensing circuitry produces a voltage across resistor R

that resembles the inductor current waveform transformed

to a voltage. If there is an increase in the power converter

load on V

OUT

, the instantaneous level of V

OUT

will drop

slightly, which will increase the voltage level on VC by

the inverting action of the voltage error amplifier. When

the increase on VC first occurs, the output of the current

averaging amplifier, VIA, will also increase momentarily

to command a larger duty cycle. This duty cycle increase

will result in a higher inductor current level, ultimately

raising the average voltage across R

. Once the average

value of the voltage on R

is equivalent to the VC level,

the voltage on VIA will revert very closely to its previous

level into the PWM and force the correct duty cycle to

maintain voltage regulation at this new higher inductor

current level. The average current amplifier is configured

as an integrator, so in steady state, the average value of

the voltage applied to its inverting input (voltage across

) will be equivalent to the voltage on its noninverting,

VC. As a result, the average value of the inductor current

is controlled in order to maintain voltage regulation. The

entire current amplifier and PWM can be simplified as a

voltage controlled current source, with the driving volt

age coming from VC. VC is commonly referred to as the

current command for this reason and the voltage on VC

is directly proportional to average inductor current, which

can prove useful for many applications.

The voltage error amplifier monitors the output voltage,

OUT

through a voltage divider and makes adjustments to

the current command as necessary to maintain regulation.

The voltage error amplifier therefore controls the outer

voltage regulation loop. The average current amplifier

makes adjustments to the inductor current as directed by

the voltage error amplifier output via VC and is commonly

referred to as the inner current-loop amplifier.

The average current mode control technique is similar to

peak current mode control except that the average current

amplifier, by virtue of its configuration as an integrator,

controls average current instead of the peak current. This

difference eliminates the peak to average current error inher

ent to peak current mode control, while maintaining most

of the advantages inherent to peak current mode control.

Average current mode control requires appropriate com

pensation for the inner current loop unlike peak current

mode control. The compensation network must have high

DC gain to minimize errors between the commanded av

erage current level and actual, high bandwidth to quickly

change the commanded current level following transient

load steps and a controlled mid-band gain to provide a

form of slope compensation unique to average current

mode control. Fortunately, the compensation components

required to ensure these sometimes conflicting require

ments have been carefully selected and are integrated

within the LTC3114-1. With the inner loop compensation

fixed internally, compensation of the outer voltage loop

as is detailed in the applications section, is similar to well

known techniques used with peak current mode control.

Inductor Current Sense and Maximum Output Current

As part of the current control loop required for current

mode control, the LTC3114-1 includes a pair of current

sensing circuits that directly measure the buck-boost

converter inductor current as shown in Figure 2. These

circuits measure the voltage dropped across switches A

and B separately and produce output currents proportional

to the switches’ voltage drop. By sensing current in this

manner, there is no additional power loss incurred, which

improves converter efficiency. The amplifier output ter

minals are summed together into a common resistor, R

connected to ground. Since switches A and B are never

conducting at the same time, the resultant waveform on R

resembles the inductor current. This replica of the induc-

tor current is used as one input to the current averaging

amplifier as described in the previous section.

e voltage error amplifier output, VC, is internally clamped

to a nominal level of 1V. Since the average inductor current

is proportional to VC, the 1V clamp level sets the maxi

mum average inductor current that can be programmed

by the inner current loop. T

aking into account the current

sense amplifier

’s gain and the value of R

, the maximum

average inductor current is approximately 1.7A (typical).

In buck mode, the output current is approximately equal

to the inductor current, I

OUT(BUCK)

≈ I

• 0.9

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The SW1/SW2 forced low time on each switching cycle

briefly disconnects the inductor from V

OUT

and V

result-

ing in slightly less output current in either buck or boost

mode for a given inductor current. In boost mode, the

output current is related to average inductor current and

duty cycle by:

OUT(BOOST)

≈ I

• (1 – D)

where D is the converter duty cycle.

Since the output current in boost mode is reduced by the

duty cycle (D), the output current rating in buck mode is

always greater than in boost mode. Also, because boost

mode operation requires a higher inductor current for a

given output current compared to buck mode, the efficiency

in boost mode will be lower due to higher I

INDUCTOR

•

DS(ON)

losses in the power switches. This will further

reduce the output current capability in boost mode. In

either operating mode, however, the inductor peak-to-peak

ripple current does not play a major role in determining the

output current capability, unlike peak current mode control.

With peak current mode control, the maximum output

current capability is reduced by the magnitude of inductor

ripple current because the peak inductor current level is the

control variable, but the average inductor current is what

determines the output current. The LTC3114 -1 measures

and controls average inductor current, and therefore, the

inductor ripple current magnitude has little effect on the

maximum current capability in contrast to an equivalent

peak current mode converter. Under most conditions in

buck mode, the LTC3114 -1 is capable of providing 1A to

the load. Under certain conditions, more output current is

possible, refer to the Typical Performance Characteristics

section for more details. In boost mode, as described

previously, the output current capability is related to the

boost ratio or duty cycle (D). For a 3.6V V

to 5V output

application, the LTC3114-1 can provide up to 500mA to

the load. Refer to the Typical Performance Characteristics

section for more detail on output current capability.

At VC levels below 135mV, the LTC3114-1 will not com

mand any current because the internal current sense signal

has a built-in

135mV

offset. Therefore, the active range of

VC is between approximately 135mV (zero current) and

1V (full current). In some applications, an external circuit

may be used to control the VC voltage level. Any such

circuit needs to have the capability to sink or source the

approximate 12µA provided by the internal error amplifier

and to pull below 135mV to disable the current command,

if necessary.

OVERLOAD CURRENT LIMIT AND ZERO CURRENT

COMPARATOR

The internal current sense waveform is also used by the

peak overload current (I

PEAK

) and zero current (I

ZERO

)

comparators. The I

PEAK

current comparator monitors I

S-

ENSE

and halts converter operation if the inductor current

level exceeds its maximum internal threshold, which is

approximately 50% above the normal maximum current

level commanded by the current control loop. An induc

tor current level of this magnitude will only occur during

a fault, such as an output short circuit or a fast V

(line)

transient. If the I

PEAK

comparator is engaged, the PWM is

halted for the remainder of the switching cycle with SW1

and SW2 held low. If V

OUT

is less than approximately 1.8V

when the peak limit occurs, then a soft-start cycle is initi-

ated. In the event that the current overload is the result

of an output short-cir

cuit condition, the

LTC3114-1 will

remain in a low frequency restart mode, keeping the on-chip

power dissipation to very low levels. If the short circuit is

removed, the LTC3114-1 will restart in the normal fashion.

The LTC3114-1 exhibits discontinuous inductor current

operation at light output loads by virtue of the I

ZERO

comparator circuit under most operating conditions. This

improves efficiency at light output loads if PWM mode

operation compared to continuous conduction mode. If

the internal current sense waveform transitions below the

internally set zero current threshold, the LTC3114-1 will

disconnect the inductor from V

OUT

, by shutting off switch

D, to prevent discharge of the output capacitor. The I

ZERO

circuitry is reset by the oscillator clock at the end of the

switching cycle. The I

ZERO

comparator threshold is set

slightly above zero current to compensate for comparator

propagation delay. In some cases, the inductor current

may reverse slightly if there is a very high voltage output

LTC3114-1

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operaTion

or small inductor resulting in a small amount of residual

energy left in the inductor following a zero current event.

In this case the LTC3114-1 SW1 waveform will display a

characteristic half sine wave between the time at which

Izero is detected and when the next switching cycle com

mences. This is because SWC is the only active (on) switch

following an

ZERO

event and this behavior is not harmful

to the LTC3114-1.

Burst Mode Operation

When the MODE pin is held low, the LTC3114-1 is config

ured for Burst Mode operation. As a result, the buck-boost

DC/DC converter will operate with normal continuous PWM

switching above a predetermined minimum output load

and will automatically transition to power saving Burst

Mode operation below this output load level. Refer to the

ypical Per

formance Characteristics section of this data

sheet to determine the Burst Mode transition threshold

for various combinations of V

and V

OUT

. If MODE is

low, at light output loads, the LTC3114-1 will go into a

standby or sleep state when the output voltage achieves

its nominal regulation level. The sleep state halts PWM

switching and powers down all non-essential functions

of the IC, significantly reducing the quiescent current

of the LTC3114-1. This greatly improves overall power

conversion efficiency when the output load is light. Since

the converter is not operating in sleep, the output volt

age will slowly decay at a rate determined by the output

load resistance and the output capacitor value. When the

output voltage has decayed by a small amount, typically

1%, the LTC3114-1 will wake and resume normal PWM

switching operation until the voltage on V

OUT

is restored to

the previous level. If the load is very light, the LTC3114-1

may only need to switch for a few cycles to restore V

OUT

and may sleep for extended periods of time, significantly

improving efficiency.

Soft-Start

The LTC3114-1 soft-start circuit minimizes input current

transients and output voltage overshoot on initial power up.

The required timing components for soft-start are internal

to the LTC3114-1 and produce a nominal soft-start dura

tion of approximately 2ms. The internal soft-start circuit

slowly

ramps

the error amplifier output, VC. In doing so,

the current command of the IC is also slowly increased,

starting from zero. It is unaffected by output loading or

output capacitor value. Soft-start is reset by undervolt

age lockout on both V

and LDO, the accurate RUN pin

comparator, thermal shutdown and the overload current

limit as described previously.

LDO REGULATOR

An internal low dropout regulator generates a nominal

4.4V rail from V

. The LDO rail powers the internal control

circuitry and power device gate drivers of the LTC3114-1.

The LDO regulator is disabled in shutdown to reduce

quiescent current and is enabled by forcing the RUN pin

above its logic threshold. The LDO regulator includes

current-limit protection to safeguard against accidental

short-circuiting of the LDO rail. In 5V V

OUT

applications,

the LDO can be driven by V

OUT

through a Schottky diode,

commonly referred to as bootstrapping. Bootstrapping can

provide a significant efficiency improvement, particularly

when V

is very high and also allows operation to the

minimum rated input voltage of 2.2V.

UNDERVOLTAGE LOCKOUT

The LTC3114-1 undervoltage lockout (UVLO) circuit dis

ables operation of the internal power switches and keeps

other IC functions in a reset state if either the input voltage

applied to V

or the LDO output voltage are below their

respective UVLO thresholds. There are two UVLO circuits,

one that monitors V

and another that monitors LDO. The

UVLO comparator has a falling voltage threshold of

2.1V (typical). If V

falls below this level, IC operation is

disabled until V

rises above 2.2V (typical), as long as

the LDO voltage is above its UVLO threshold. The LDO

UVLO has a falling voltage threshold of 2.4V (typical). If

the LDO voltage falls below this threshold, IC operation

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Manufacturer:

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Description:

Switching Voltage Regulators 40V, 1A Sync. Buck-Boost Converter with Programmable Current Limit

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LTC3114HDHC-1#TRPBF

LTC3114HDHC-1#TRPBF

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