OP275GSZ Datasheet OP275GSZ-REEL, OP275GSZ-REEL7, OP275GSZ | P10-P12

OP275

–8–

OP275

–9–

+15V

0.1

F

OUT

2k

–15V

10F

0.1

F

1/2

OP275

10F

–

Figure 12. Unity Gain Follower

0.1



+15V



OUT



–15V



0.1



10pF

4.99k



2.49k



4.99k



1/2

OP275

–

Figure 13. Unity Gain Inverter

In inverting and noninverting applications, the feedback resis-

tance forms a pole with the source resistance and capacitance

and C

) and the OP275’s input capacitance (C

), as shown

in Figure 14. With R

and R

in the kilohm range, this pole

can create excess phase shift and even oscillation. A small

capacitor, C

, in parallel and R

eliminates this problem.

By setting R

+ C

) = R

, the effect of the feedback

pole is completely removed.

16V–20V

0.1F

V+

5V

1k

D1 D2

+15V

2N4416

1k

OUTPUT

(TO SCOPE)

1F

10k

2k

750

2N2222A

15k

–15V

1N4148

DUT

1/2 OP260AJ

16V–20V

0.1F

10k

– +

–

SCHOTTKY DIODES D1–D4 AR

HEWLETT-PACKARD HP5082-2835

IC1 IS 1/2 OP260AJ

IC2 IS PMI OP41EJ

V–

222

–

Figure 11. OP275’s Settling Time Test Fixture

OUT

–

Figure 14. Compensating the Feedback Pole

Attention to Source Impedances Minimizes Distortion

Since the OP275 is a very low distortion amplier, careful atten-

tion should be given to source impedances seen by both inputs.

As with many FET-type ampliers, the p-channel JFETs in the

OP275’s input stage exhibit a gate-to-source capacitance that var-

ies with the applied input voltage. In an inverting conguration,

the inverting input is held at a virtual ground and, as such, does

not vary with input voltage. Thus, since the gate-to-source voltage

is constant, there is no distortion due to input capacitance modu-

lation. In noninverting applications, however, the gate-to-source

voltage is not constant. The resulting capacitance modulation

can cause distortion above 1 kHz if the input impedance is

greater than 2 k



and unbalanced.

OP275

OUT

= R

//R

IF R

//R

> 2k



FOR MINIMUM DISTORTION

–

Figure 15. Balanced Input Impedance to Minimize

Distortion in Noninverting Amplier Circuits

Figure 15 shows some guidelines for maximizing the distortion

performance of the OP275 in noninverting applications. The best

way to prevent unwanted distortion is to ensure that the parallel

combination of the feedback and gain setting resistors (R

and

) is less than 2 k



. Keeping the values of these resistors small

has the added benets of reducing the thermal noise of the circuit

REV. C

OP275

–10–

OP275

–11–

and dc offset errors. If the parallel combination of R

and R

larger than 2 k



, then an additional resistor, R

, should be used

in series with the noninverting input. The value of R

is deter-

mined by the parallel combination of R

and R

to maintain the

low distortion performance of the OP275.

Driving Capacitive Loads

The OP275 was designed to drive both resistive loads to 600



and capacitive loads of over 1000 pF and maintain stability. While

there is a degradation in bandwidth when driving capacitive loads,

the designer need not worry about device stability. The graph in

Figure 16 shows the 0 dB bandwidth of the OP275 with capaci-

tive loads from 10 pF to 1000 pF.

0 200 400 600 800 1000

LOAD

– pF

BANDWIDTH – MHz

Figure 16. Bandwidth vs. C

LOAD

High Speed, Low Noise Differential Line Driver

The circuit in Figure 17 is a unique line driver widely used in

industrial applications. With ±18 V supplies, the line driver can

deliver a differential signal of 30 V p-p into a 2.5 k



load. The

high slew rate and wide bandwidth of the OP275 combine to

yield a full power bandwidth of 130 kHz while the low noise

front end produces a referred-to-input noise voltage spectral

density of 10 nV/



Hz.



R11



10k



R12



R10



– V

= V

A1 = 1/2 OP275

A2, A3 = 1/2 OP275

GAIN =

SET R2, R4, R5 = R1 AND R6, R7, R8 = R3



–

Figure 17. High Speed, Low Noise Differential Line Driver

The design is a transformerless, balanced transmission system

where output common-mode rejection of noise is of paramount

importance. Like the transformer based design, either output can

be shorted to ground for unbalanced line driver applications

without changing the circuit gain of 1. Other circuit gains can be

set according to the equation in the diagram. This allows the

design to be easily set to noninverting, inverting, or differential

operation.

A 3-Pole, 40 kHz Low-Pass Filter

The closely matched and uniform ac characteristics of the OP275

make it ideal for use in GIC (Generalized Impedance Converter)

and FDNR (Frequency-Dependent Negative Resistor) lter

applications. The circuit in Figure 18 illustrates a linear-phase,

3-pole, 40 kHz low-pass lter using an OP275 as an inductance

simulator (gyrator). The circuit uses one OP275 (A2 and A3) for

the FDNR and one OP275 (A1 and A4) as an input buffer and

bias current source for A3. Amplier A4 is congured in a gain

of 2 to set the pass band magnitude response to 0 dB. The ben-

ets of this lter topology over classical approaches are that the

op amp used in the FDNR is not in the signal path and that the

lter’s performance is relatively insensitive to component varia-

tions. Also, the conguration is such that large signal levels can

be handled without overloading any of the lter’s internal nodes.

As shown in Figure 19, the OP275’s symmetric slew rate and low

distortion produce a clean, well behaved transient response.

95.3k

787

2200pF

1.82k

2200pF

1.87k

1.82k

4.12k

2200pF

100k

1k

OUT

A1, A4 = 1/2 OP275

A2, A3 = 1/2 OP275

–

Figure 18. A 3-Pole, 40 kHz Low-Pass Filter

OUT

10V p-p

10kHz

SCALE: VERTICAL–2V/ DIV

HORIZONTAL–10

s/ DIV

100

Figure 19. Low-Pass Filter Transient Response

REV. C

OP275

–10–

OP275

–11–

OP275 SPICE Model

* Node assignments

* noninverting input

* inverting input

* positive supply

* negative supply

* output

.SUBCKT OP275 1 2 99 50 34

* INPUT STAGE & POLE AT 100 MHz

R3 5 51 2.188

R4 6 51 2.188

CIN 1 2 3.7E-12

CM1 1 98 7.5E-12

CM2 2 98 7.5E-12

C2 5 6 364E-12

I1 97 4 100E-3

IOS 1 2 1E-9

EOS 9 3 POLY(1) 26 28 0.5E-3 1

Q1 5 2 7 QX

Q2 6 9 8 QX

R5 7 4 1.672

R6 8 4 1.672

D1 2 36 DZ

D2 1 36 DZ

EN 3 1 10 0 1

GN1 0 2 13 0 1E-3

GN2 0 1 16 0 1E-3

EREF 98 0 28 0 1

EP 97 0 99 0 1

EM 51 0 50 0 1

* VOLTAGE NOISE SOURCE

DN1 35 10 DEN

DN2 10 11 DEN

VN1 35 0 DC 2

VN2 0 11 DC 2

* CURRENT NOISE SOURCE

DN3 12 13 DIN

DN4 13 14 DIN

VN3 12 0 DC 2

VN4 0 14 DC 2

* CURRENT NOISE SOURCE

DN5 15 16 DIN

DN6 16 17 DIN

VN5 15 0 DC 2

VN6 0 17 DC 2

* GAIN STAGE & DOMINANT POLE AT 32 Hz

R7 18 98 1.09E6

C3 18 98 4.55E-9

G1 98 18 5 6 4.57E-1

V2 97 19 1.35

V3 20 51 1.35

D3 18 19 DX

D4 20 18 DX

* POLE/ZERO PAIR AT 1.5 MHz/2.7 MHz

R8 21 98 1E-3

R9 21 22 1.25E-3

C4 22 98 47.2E-12

G2 98 21 18 28 1E-3

* POLE AT 100 MHz

R10 23 98 1

C5 23 98 1.59E-9

G3 98 23 21 28 1

* POLE AT 100 MHz

R11 24 98 1

C6 24 98 1.59E-9

G4 98 24 23 28 1

* COMMON-MODE GAIN NETWORK WITH ZERO AT

1 kHz

R12 25 26 1E6

C7 25 26 1.5915E-12

R13 26 98 1

E2 25 98 POLY(2) 1 98 2 98 0 2.50

2.50

* POLE AT 100 MHz

R14 27 98 1

C8 27 98 1.59E-9

G5 98 27 24 28 1

* OUTPUT STAGE

R15 28 99 100E3

R16 28 50 100E3

C9 28 50 1E-6

ISY 99 50 1.85E-3

R17 29 99 100

R18 29 50 100

L2 29 34 1E-9

G6 32 50 27 29 10E-3

G7 33 50 29 27 10E-3

G8 29 99 99 27 10E-3

G9 50 29 27 50 10E-3

V4 30 29 1.3

V5 29 31 3.8

F1 29 0 V4 1

F2 0 29 V5 1

D5 27 30 DX

D6 31 27 DX

D7 99 32 DX

D8 99 33 DX

D9 50 32 DY

D10 50 33 DY

* MODELS USED

.MODEL QX PNP(BF=5E5)

.MODEL DX D(IS=1E-12)

.MODEL DY D(IS=1E-15 BV=50)

.MODEL DZ D(IS=1E-15 BV=7.0)

.MODEL DEN D(IS=1E-12 RS=4.35K KF=1.95E-15

AF=1)

.MODEL DIN D(IS=1E-12 RS=268 KF=1.08E-15 AF=1)

.ENDS

REV. C

P1-P3 P4-P6 P7-P9 P10-P12 P13-P13

OP275GSZ

Mfr. #:

Buy OP275GSZ

Manufacturer:

Analog Devices Inc.

Description:

Audio Amplifiers Bipolar/JFET Audio Dual 1mV 2nA

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OP275GSZ

OP275GSZ

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