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LTC1872ES6#TRMPBF

P1-P3P4-P6P7-P9P10-P12P13-P14
LTC1872
7
1872fa
For more information www.linear.com/LTC1872
applicaTions inForMaTion
Inductor Selection
When selecting the inductor, keep in mind that inductor
saturation current has to be greater than the current limit
set by the current sense resistor. Also, keep in mind that
the DC resistance of the inductor will affect the efficiency.
Off the shelf inductors are available from Murata, Coilcraft,
Toko, Panasonic, Coiltronics and many other suppliers.
Power MOSFET Selection
The main selection criteria for the power MOSFET are the
threshold voltage V
GS(TH)
, the “on” resistance R
DS(ON)
,
reverse transfer capacitance C
RSS
and total gate charge.
Since the LTC1872 is designed for operation down to low
input voltages, a logic level threshold MOSFET (R
DS(ON)
guaranteed at V
GS
= 2.5V) is required for applications
that work close to this voltage. When these MOSFETs are
used, make sure that the input supply to the LTC1872 is
less than the absolute maximum V
GS
rating, typically 8V.
The required minimum R
DS(ON)
of the MOSFET is governed
by its allowable power dissipation given by:
R
DS(ON)
≅
P
P
DC
( )
I
IN
2
1+δp
( )
where P
P
is the allowable power dissipation and δp is the
temperature dependency of R
DS(ON)
. (1 + δp) is generally
given for a MOSFET in the form of a normalized R
DS(ON)
vs temperature curve, but δp = 0.005/°C can be used as
an approximation for low voltage MOSFETs. DC is the
maximum operating duty cycle of the LTC1872.
Output Diode Selection
Under normal load conditions, the average current con
-
ducted by the diode in a boost converter is equal to the
output load current:
D(avg)
=
OUT
It is important to adequately specify the diode peak cur-
rent and average power dissipation so as not to exceed
the diode ratings.
Schottky diodes are recommended for low for
ward drop
and fast switching times. Remember to keep lead length
short and observe proper grounding (see Board Layout
Checklist) to avoid ringing and increased dissipation.
C
IN
and C
OUT
Selection
To prevent large input voltage ripple, a low ESR input
capacitor sized for the maximum RMS current must be
used. The maximum RMS capacitor current for a boost
converter is approximately equal to:
C
IN
Required I
RMS
≈
0.3
( )
I
RIPPLE
where I
RIPPLE
is as defined in the Inductor Value Calcula-
tion section.
Note that capacitor manufacturer’s ripple current ratings are
often based on 2000 hours of life. This makes it advisable to
further derate the capacitor, or to choose a capacitor rated
at a higher temperature than required. Several capacitors
may be paralleled to meet the size or height requirements
in the design. Due to the high operating frequency of the
LTC1872, ceramic capacitors can also be used for C
IN
.
Always consult the manufacturer if there is any question.
The selection of C
OUT
is driven by the required effective
series resistance (ESR). Typically, once the ESR require-
ment is satisfied, the capacitance is adequate for filtering.
The output ripple (∆V
OUT
) is approximated by:
ΔV
OUT
≈ I
O
•
V
OUT
+ V
D
V
IN
+
I
RIPPLE
2
•
ESR
2
+
1
2πfC
OUT
2
1
2
LTC1872
8
1872fa
For more information www.linear.com/LTC1872
applicaTions inForMaTion
where f is the operating frequency, C
OUT
is the output
capacitance and I
RIPPLE
is the ripple current in the inductor.
Manufacturers such as Nichicon, United Chemicon and
Sanyo should be considered for high performance through-
hole capacitors. The OS-CON semiconductor dielectric
capacitor available from Sanyo has the lowest ESR (size)
product of any aluminum electrolytic at a somewhat higher
price. The output capacitor RMS current is approximately
equal to:
I
PK
• DC−DC
2
where I
PK
is the peak inductor current and DC is the switch
duty cycle.
When using electrolytic output capacitors, if the ripple and
ESR requirements are met, there is likely to be far more
capacitance than required.
In surface mount applications, multiple capacitors may
have to be paralleled to meet the ESR or RMS current
handling requirements of the application. Aluminum
electrolytic and dry tantalum capacitors are both available
in surface mount configurations. An excellent choice of
tantalum capacitors is the AVX TPS and KEMET T510
series of surface mount tantalum capacitors. Also,
ceramic capacitors in X5R pr X7R dielectrics offer excel-
lent performance.
Low Supply Operation
Although the LTC1872 can function down to approxi-
mately 2.0V, the maximum allowable output current is
reduced when
V
IN
decreases below 3V. Figure 3 shows
the amount of change as the supply is reduced down to
2V. Also shown in Figure 3 is the effect of V
IN
on V
REF
as
V
IN
goes below 2.3V.
Setting Output Voltage
The LTC1872 develops a 0.8V reference voltage between
the feedback (Pin 3) terminal and ground (see Figure 4).
By selecting resistor R1, a constant current is caused to
flow through R1 and R2 to set the overall output voltage.
The regulated output voltage is determined by:
V
OUT
= 0.8V 1+
R2
R1
INPUT VOLTAGE (V)
2.0
NORMALIZED VOLTAGE (%)
105
100
95
90
85
80
75
2.2 2.4 2.6 2.8
1872 F03
3.0
V
REF
V
ITH
Figure 4. Setting Output Voltage
Figure 3. Line Regulation of V
REF
and V
ITH
3
V
FB
V
OUT
LTC1872
R1
1872 F04
R2
LTC1872
9
1872fa
For more information www.linear.com/LTC1872
applicaTions inForMaTion
For most applications, an 80k resistor is suggested for
R1. To prevent stray pickup, locate resistors R1 and R2
close to LTC1872.
Efficiency Considerations
The efficiency of a switching regulator is equal to the output
power divided by the input power times 100%. It is often
useful to analyze individual losses to determine what is
limiting the efficiency and which change would produce
the most improvement. Efficiency can be expressed as:
Efficiency = 100% – (η1 + η2 + η3 + ...)
where η1, η2, etc. are the individual losses as a percent-
age of input power.
Although all
dissipative elements in the circuit produce
losses, four main sources usually account for most of the
losses in LTC1872 circuits: 1) LTC1872 DC bias current, 2)
MOSFET gate charge current, 3) I
2
R losses and 4) voltage
drop of the output diode.
1. The V
IN
current is the DC supply current, given in the
electrical characteristics, that excludes MOSFET driver
and control currents. V
IN
current results in a small loss
which increases with V
IN
.
2. MOSFET gate charge current results from switching
the gate capacitance of the power MOSFET. Each
time a MOSFET gate is switched from low to high to
low again, a packet of charge, dQ, moves from V
IN
to ground. The resulting dQ/dt is a current out of V
IN
which is typically much larger than the contoller’s DC
supply current. In continuous mode, I
GATECHG
= f(Qp).
3. I
2
R losses are predicted from the DC resistances of
the MOSFET, inductor and current sense resistor.
The MOSFET R
DS(ON)
multiplied by duty cycle times
the average output current squared can be summed
with I
2
R losses in the inductor ESR in series with the
current sense resistor.
4. The output diode is a major source of power loss at high
currents. The diode loss is calculated by multiplying
the forward voltage by the load current.
5. Transition losses apply to the external MOSFET and
increase at higher operating frequencies and input
voltages. Transition losses can be estimated from:
Transition Loss = 2(V
IN
)
2
I
IN(MAX)
C
RSS
(f)
Other losses, including C
IN
and C
OUT
ESR dissipative
losses, and inductor core losses, generally account for
less than 2% total additional loss.
P1-P3P4-P6P7-P9P10-P12P13-P14

LTC1872ES6#TRMPBF

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Analog Devices / Linear Technology
Description:
Switching Voltage Regulators SOT-23 Boost Controller
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