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HIP6302CBZA-T

P1-P3P4-P6P7-P9P10-P12P13-P15P16-P16
13
FN4766.3
December 27, 2004
As discussed previously, the voltage drop across each Q2
transistor at the point in time when current is sampled is
r
DSON
(Q2) x I
SAMPLE
. The voltage at Q2’s drain, the
PHASE node, is applied through the R
ISEN
resistor to the
HIP6302 ISEN pin. This pin is held at virtual ground, so the
current into ISEN is:
I
SENSE
= I
SAMPLE
x r
DS(ON)
(Q2) / R
ISEN
.
R
Isen
= I
SAMPLE
x r
DS(ON)
(Q2) / 50µA
Example: From the previous conditions,
where I
LT
= 50A,
I
SAMPLE
= 25.49A,
r
DS(ON)
(Q2) = 4mΩ
Then: R
ISEN
= 2.04K and
I
CURRENT TRIP
= 165%
Short circuit I
LT
= 82.5A.
Channel Frequency Oscillator
The channel oscillator frequency is set by placing a resistor,
R
T
, to ground from the FS/DIS pin. Figure 10 is a curve
showing the relationship between frequency, F
SW,
and
resistor R
T
. To avoid pickup by the FS/DIS pin, it is important
to place this resistor next to the pin. If this pin is also used to
disable the converter, it is also important to locate the pull-
down device next to this pin.
Layout Considerations
MOSFETs switch very fast and efficiently. The speed with
which the current transitions from one device to another
causes voltage spikes across the interconnecting
impedances and parasitic circuit elements. These voltage
spikes can degrade efficiency, radiate noise into the circuit
and lead to device over-voltage stress. Careful component
layout and printed circuit design minimizes the voltage
spikes in the converter. Consider, as an example, the turnoff
transition of the upper PWM MOSFET. Prior to turnoff, the
upper MOSFET was carrying channel current. During the
turnoff, current stops flowing in the upper MOSFET and is
picked up by the lower MOSFET. Any inductance in the
switched current path generates a large voltage spike during
the switching interval. Careful component selection, tight
layout of the critical components, and short, wide circuit
traces minimize the magnitude of voltage spikes. Contact
Intersil for evaluation board drawings of the component
placement and printed circuit board.
There are two sets of critical components in a DC-DC
converter using a HIP6302 controller and a HIP6601 gate
driver. The power components are the most critical because
they switch large amounts of energy. Next are small signal
components that connect to sensitive nodes or supply critical
bypassing current and signal coupling.
The power components should be placed first. Locate the
input capacitors close to the power switches. Minimize the
length of the connections between the input capacitors, C
IN
,
and the power switches. Locate the output inductors and
output capacitors between the MOSFETs and the load.
Locate the gate driver close to the MOSFETs.
The critical small components include the bypass capacitors
for VCC and PVCC on the gate driver ICs. Locate the bypass
capacitor, C
BP
, for the HIP6302 controller close to the
device. It is especially important to locate the resistors
associated with the input to the amplifiers close to their
respective pins, since they represent the input to feedback
amplifiers. Resistor R
T
, that sets the oscillator frequency
should also be located next to the associated pin. It is
especially important to place the R
SEN
resistor(s) at the
respective terminals of the HIP6302.
A multi-layer printed circuit board is recommended. Figure 11
shows the connections of the critical components for one
output channel of the converter. Note that capacitors C
IN
and
C
OUT
could each represent numerous physical capacitors.
Dedicate one solid layer, usually the middle layer of the PC
board, for a ground plane and make all critical component
ground connections with vias to this layer. Dedicate another
solid layer as a power plane and break this plane into smaller
islands of common voltage levels. Keep the metal runs from
the PHASE terminal to inductor L
O1
short. The power plane
should support the input power and output power nodes. Use
copper filled polygons on the top and bottom circuit layers for
the phase nodes. Use the remaining printed circuit layers for
small signal wiring. The wiring traces from the driver IC to the
MOSFET gate and source should be sized to carry at least
one ampere of current.
50 10010 20 200 500 1,000 5,000 10,0002,000
1
2
5
10
20
50
100
200
500
1,000
R
T
(kΩ)
CHANNEL OSCILLATOR FREQUENCY, F
SW
(kHz)
FIGURE 10. RESISTANCE R
T
vs FREQUENCY
HIP6302HIP6302
14
FN4766.3
December 27, 2004
Component Selection Guidelines
Output Capacitor Selection
The output capacitor is selected to meet both the dynamic
load requirements and the voltage ripple requirements. The
load transient for the microprocessor CORE is characterized
by high slew rate (di/dt) current demands. In general,
multiple high quality capacitors of different size and dielectric
are paralleled to meet the design constraints.
Modern microprocessors produce severe transient load rates.
High frequency capacitors supply the initially transient current
and slow the load rate-of-change seen by the bulk capacitors.
The bulk filter capacitor values are generally determined by
the ESR (effective series resistance) and voltage rating
requirements rather than actual capacitance requirements.
High frequency decoupling capacitors should be placed as
close to the power pins of the load as physically possible. Be
careful not to add inductance in the circuit board wiring that
could cancel the usefulness of these low inductance
components. Consult with the manufacturer of the load on
specific decoupling requirements.
Use only specialized low-ESR capacitors intended for
switching-regulator applications for the bulk capacitors. The
bulk capacitor’s ESR determines the output ripple voltage
and the initial voltage drop following a high slew-rate
transient’s edge. In most cases, multiple capacitors of small
case size perform better than a single large case capacitor.
Bulk capacitor choices include aluminum electrolytic, OS-
Con, Tantalum and even ceramic dielectrics. An aluminum
electrolytic capacitor’s ESR value is related to the case size
with lower ESR available in larger case sizes. However, the
equivalent series inductance (ESL) of these capacitors
increases with case size and can reduce the usefulness of
the capacitor to high slew-rate transient loading.
Unfortunately, ESL is not a specified parameter. Consult the
capacitor manufacturer and measure the capacitor’s
impedance with frequency to select a suitable component.
Output Inductor Selection
One of the parameters limiting the converter’s response to a
load transient is the time required to change the inductor
current. Small inductors in a multi-phase converter reduces
the response time without significant increases in total ripple
current.
The output inductor of each power channel controls the
ripple current. The control IC is stable for channel ripple
current (peak-to-peak) up to twice the average current. A
single channel’s ripple current is approximately:
The current from multiple channels tend to cancel each other
and reduce the total ripple current. Figure 12 gives the total
ripple current as a function of duty cycle, normalized to the
parameter at zero duty cycle. To determine the
total ripple current from the number of channels and the duty
cycle, multiply the y-axis value by .
Small values of output inductance can cause excessive power
dissipation. The HIP6303 is designed for stable operation for
ripple currents up to twice the load current. However, for this
condition, the RMS current is 115% above the value shown in
the following MOSFET Selection and Considerations section.
With all else fixed, decreasing the inductance could increase
the power dissipated in the MOSFETs by 30%.
∆I
V
IN
V
OUT
–
F
SW
xL
--------------------------------
V
OUT
V
IN
----------------
×=
V
CORE
+12V
VIA CONNECTION TO GROUND PLANE
ISLAND ON POWER PLANE LAYER
ISLAND ON CIRCUIT PLANE LAYER
L
O1
C
OUT
C
IN
+5V
IN
KEY
PHASE
VCC
USE INDIVIDUAL METAL RUNS
COMP
HIP6302
PWM
R
T
R
IN
R
FB
C
BP
FB
VSEN
ISEN
R
SEN
HIP6601
C
BOOT
C
BP
C
T
V
CC
FS/DIS
PVCC
LOCATE NEXT TO IC PIN
LOCATE NEXT
TO FB PIN
LOCATE NEXT TO IC PIN(S)
ISOLATE OUTPUT STAGES
FOR EACH CHANNEL TO HELP
LOCATE NEAR TRANSISTOR
FIGURE 11. PRINTED CIRCUIT BOARD POWER PLANES AND ISLANDS
Vo()LF
S
⋅()⁄
Vo()LxF
SW
()⁄
HIP6302HIP6302
15
FN4766.3
December 27, 2004
Input Capacitor Selection
The important parameters for the bulk input capacitors are
the voltage rating and the RMS current rating. For reliable
operation, select bulk input capacitors with voltage and
current ratings above the maximum input voltage and
largest RMS current required by the circuit. The capacitor
voltage rating should be at least 1.25 times greater than the
maximum input voltage and a voltage rating of 1.5 times is
a conservative guideline. The RMS current required for a
multi-phase converter can be approximated with the aid of
Figure 13.
First determine the operating duty ratio as the ratio of the
output voltage divided by the input voltage. Find the Current
Multiplier from the curve with the appropriate power
channels. Multiply the current multiplier by the full load
output current. The resulting value is the RMS current rating
required by the input capacitor.
Use a mix of input bypass capacitors to control the voltage
overshoot across the MOSFETs. Use ceramic capacitance
for the high frequency decoupling and bulk capacitors to
supply the RMS current. Small ceramic capacitors should
be placed very close to the drain of the upper MOSFET to
suppress the voltage induced in the parasitic circuit
impedances.
For bulk capacitance, several electrolytic capacitors (Panasonic
HFQ series or Nichicon PL series or Sanyo MV-GX or
equivalent) may be needed. For surface mount designs, solid
tantalum capacitors can be used, but caution must be exercised
with regard to the capacitor surge current rating. These
capacitors must be capable of handling the surge-current at
power-up. The TPS series available from AVX, and the 593D
series from Sprague are both surge current tested.
MOSFET Selection and Considerations
In high-current PWM applications, the MOSFET power
dissipation, package selection and heatsink are the
dominant design factors. The power dissipation includes two
loss components; conduction loss and switching loss. These
losses are distributed between the upper and lower
MOSFETs according to duty factor (see the following
equations). The conduction losses are the main component
of power dissipation for the lower MOSFETs, Q2 and Q4 of
Figure 1. Only the upper MOSFETs, Q1 and Q3 have
significant switching losses, since the lower device turns on
and off into near zero voltage.
The equations assume linear voltage-current transitions and
do not model power loss due to the reverse-recovery of the
lower MOSFETs body diode. The gate-charge losses are
dissipated by the Driver IC and don't heat the MOSFETs.
However, large gate-charge increases the switching time,
t
SW
which increases the upper MOSFET switching losses.
Ensure that both MOSFETs are within their maximum
junction temperature at high ambient temperature by
calculating the temperature rise according to package
thermal-resistance specifications. A separate heatsink may
be necessary depending upon MOSFET power, package
type, ambient temperature and air flow.
A diode, anode to ground, may be placed across Q2 and Q4
of Figure 1. These diodes function as a clamp that catches
the negative inductor swing during the dead time between
the turn off of the lower MOSFETs and the turn on of the
upper MOSFETs. The diodes must be a Schottky type to
prevent the lossy parasitic MOSFET body diode from
conducting. It is usually acceptable to omit the diodes and let
the body diodes of the lower MOSFETs clamp the negative
inductor swing, but efficiency could drop one or two percent
as a result. The diode's rated reverse breakdown voltage
must be greater than the maximum input voltage.
1.0
0.8
0.6
0.4
0.2
0
0
0.1 0.2 0.3 0.4 0.5
DUTY CYCLE (V
O
/V
IN
)
RIPPLE CURRENT (A
PEAK-PEAK
)
V
O
/ (L
X
F
SW
)
SINGLE
CHANNEL
2 CHANNEL
3 CHANNEL
4 CHANNEL
FIGURE 12. RIPPLE CURRENT vs DUTY CYCLE
0.5
0.4
0.3
0.2
0.1
0
0
0.1 0.2 0.3 0.4 0.5
DUTY CYCLE (V
O
/V
IN
)
CURRENT MULTIPLIER
SINGLE
CHANNEL
3 CHANNEL
4 CHANNEL
2 CHANNEL
FIGURE 13. CURRENT MULTIPLIER vs DUTY CYCLE
P
UPPER
I
O
2
r
DS ON()
× V
OUT
×
V
IN
------------------------------------------------------------
I
O
V
IN
× t
SW
× F
SW
×
2
----------------------------------------------------------+=
P
LOWER
I
O
2
r×
DS ON()
V
IN
V
OUT
–()×
V
IN
---------------------------------------------------------------------------------=
HIP6302HIP6302
P1-P3P4-P6P7-P9P10-P12P13-P15P16-P16

HIP6302CBZA-T

Mfr. #:
Buy HIP6302CBZA-T
Manufacturer:
Renesas / Intersil
Description:
Switching Controllers TWO PHS SYNCH BUCK CNTRLR VRM 9 X
Lifecycle:
New from this manufacturer.
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