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MIC4424YM

P1-P3P4-P6P7-P9P10-P12P13-P13
July 2005 7 MIC4423/4424/4425
MIC4423/4424/4425 Micrel, Inc.
Application Information
Although the MIC4423/24/25 drivers have been specifically
constructed to operate reliably under any practical circum-
stances, there are nonetheless details of usage which will
provide better operation of the device.
Supply Bypassing
Charging and discharging large capacitive loads quickly
requires large currents. For example, charging 2000pF from
0 to 15 volts in 20ns requires a constant current of 1.5A. In
practice, the charging current is not constant, and will usually
peak at around 3A. In order to charge the capacitor, the driver
must be capable of drawing this much current, this quickly,
from the system power supply. In turn, this means that as far
as the driver is concerned, the system power supply, as seen
by the driver, must have a VERY low impedance.
As a practical matter, this means that the power supply bus
must be capacitively bypassed at the driver with at least
100X the load capacitance in order to achieve optimum
driving speed. It also implies that the bypassing capacitor
must have very low internal inductance and resistance at
all frequencies of interest. Generally, this means using two
capacitors, one a high-performance low ESR film, the other
a low internal resistance ceramic, as together the valleys in
their two impedance curves allow adequate performance over
a broad enough band to get the job done. PLEASE NOTE
that many film capacitors can be sufficiently inductive as to
be useless for this service. Likewise, many multilayer ceramic
capacitors have unacceptably high internal resistance. Use
capacitors intended for high pulse current service (in-house
we use WIMA™ film capacitors and AVX Ramguard™ ceram-
ics; several other manufacturers of equivalent devices also
exist). The high pulse current demands of capacitive drivers
also mean that the bypass capacitors must be mounted
very close to the driver in order to prevent the effects of lead
inductance or PCB land inductance from nullifying what you
are trying to accomplish. For optimum results the sum of the
lengths of the leads and the lands from the capacitor body to
the driver body should total 2.5cm or less.
Bypass capacitance, and its close mounting to the driver serves
two purposes. Not only does it allow optimum performance
from the driver, it minimizes the amount of lead length radiat-
ing at high frequency during switching, (due to the large Δ I)
thus minimizing the amount of EMI later available for system
disruption and subsequent cleanup. It should also be noted
that the actual frequency of the EMI produced by a driver is
not the clock frequency at which it is driven, but is related to
the highest rate of change of current produced during switch-
ing, a frequency generally one or two orders of magnitude
higher, and thus more difficult to filter if you let it permeate your
system. Good bypassing practice is essential to proper
operation of high speed driver ICs.
Grounding
Both proper bypassing and proper grounding are necessary
for optimum driver operation. Bypassing capacitance only
allows a driver to turn the load ON. Eventually (except in rare
circumstances) it is also necessary to turn the load OFF. This
requires attention to the ground path. Two things other than
the driver affect the rate at which it is possible to turn a load
off: The adequacy of the grounding available for the driver,
and the inductance of the leads from the driver to the load.
The latter will be discussed in a separate section.
Best practice for a ground path is obviously a well laid out
ground plane. However, this is not always practical, and a
poorly-laid out ground plane can be worse than none. Attention
to the paths taken by return currents even in a ground plane
is essential. In general, the leads from the driver to its load,
the driver to the power supply, and the driver to whatever is
driving it should all be as low in resistance and inductance
as possible. Of the three paths, the ground lead from the
driver to the logic driving it is most sensitive to resistance or
inductance, and ground current from the load are what is most
likely to cause disruption. Thus, these ground paths should
be arranged so that they never share a land, or do so for as
short a distance as is practical.
To illustrate what can happen, consider the following: The
inductance of a 2cm long land, 1.59mm (0.062") wide on a
PCB with no ground plane is approximately 45nH. Assum-
ing a dl/dt of 0.3A/ns (which will allow a current of 3A to flow
after 10ns, and is thus slightly slow for our purposes) a volt-
age of 13.5 Volts will develop along this land in response to
our postulated ∆Ι. For a 1cm land, (approximately 15nH) 4.5
Volts is developed. Either way, anyone using TTL-level input
signals to the driver will find that the response of their driver
has been seriously degraded by a common ground path for
input to and output from the driver of the given dimensions.
Note that this is before accounting for any resistive drops in
the circuit. The resistive drop in a 1.59mm (0.062") land of
2oz. Copper carrying 3A will be about 4mV/cm (10mV/in) at
DC, and the resistance will increase with frequency as skin
effect comes into play.
The problem is most obvious in inverting drivers where the
input and output currents are in phase so that any attempt
to raise the driver’s input voltage (in order to turn the driver’s
load off) is countered by the voltage developed on the com-
mon ground path as the driver attempts to do what it was
supposed to. It takes very little common ground path, under
these circumstances, to alter circuit operation drastically.
Output Lead Inductance
The same descriptions just given for PCB land inductance
apply equally well for the output leads from a driver to its load,
except that commonly the load is located much further away
from the driver than the driver’s ground bus.
Generally, the best way to treat the output lead inductance
problem, when distances greater than 4cm (2") are involved,
requires treating the output leads as a transmission line. Un-
fortunately, as both the output impedance of the driver and the
input impedance of the MOSFET gate are at least an order of
magnitude lower than the impedance of common coax, using
coax is seldom a cost-effective solution. A twisted pair works
about as well, is generally lower in cost, and allows use of a
wider variety of connectors. The second wire of the twisted
pair should carry common from as close as possible to the
MIC4423/4424/4425 Micrel, Inc.
MIC4423/4424/4425 8 July 2005
ground pin of the driver directly to the ground terminal of the
load. Do not use a twisted pair where the second wire in the
pair is the output of the other driver, as this will not provide a
complete current path for either driver. Likewise, do not use
a twisted triad with two outputs and a common return unless
both of the loads to be driver are mounted extremely close
to each other, and you can guarantee that they will never be
switching at the same time.
For output leads on a printed circuit, the general rule is to make
them as short and as wide as possible. The lands should also
be treated as transmission lines: i.e. minimize sharp bends,
or narrowings in the land, as these will cause ringing. For a
rough estimate, on a 1.59mm (0.062") thick G-10 PCB a pair
of opposing lands each 2.36mm (0.093") wide translates to a
characteristic impedance of about 50Ω. Half that width suffices
on a 0.787mm (0.031") thick board. For accurate impedance
matching with a MIC4423/24/25 driver, on a 1.59mm (0.062")
board a land width of 42.75mm (1.683") would be required,
due to the low impedance of the driver and (usually) its load.
This is obviously impractical under most circumstances.
Generally the tradeoff point between lands and wires comes
when lands narrower than 3.18mm (0.125") would be required
on a 1.59mm (0.062") board.
To obtain minimum delay between the driver and the load, it
is considered best to locate the driver as close as possible to
the load (using adequate bypassing). Using matching trans-
formers at both ends of a piece of coax, or several matched
lengths of coax between the driver and the load, works in
theory, but is not optimum.
Driving at Controlled Rates
Occasionally there are situations where a controlled rise or
fall time (which may be considerably longer than the normal
rise or fall time of the driver’s output) is desired for a load. In
such cases it is still prudent to employ best possible practice
in terms of bypassing, grounding and PCB layout, and then
reduce the switching speed of the load (NOT the driver) by
adding a noninductive series resistor of appropriate value
between the output of the driver and the load. For situations
where only rise or only fall should be slowed, the resistor can
be paralleled with a fast diode so that switching in the other
direction remains fast. Due to the Schmitt-trigger action of the
driver’s input it is not possible to slow the rate of rise (or fall)
of the driver’s input signal to achieve slowing of the output.
Input Stage
The input stage of the MIC4423/24/25 consists of a single-
MOSFET class A stage with an input capacitance of ≤38pF.
This capacitance represents the maximum load from the
driver that will be seen by its controlling logic. The drain load
on the input MOSFET is a –2mA current source. Thus, the
quiescent current drawn by the driver varies, depending on
the logic state of the input.
Following the input stage is a buffer stage which provides
~400mV of hysteresis for the input, to prevent oscillations
when slowly-changing input signals are used or when noise
is present on the input. Input voltage switching threshold is
approximately 1.5V which makes the driver directly compat-
ible with TTL signals, or with CMOS powered from any supply
voltage between 3V and 15V.
The MIC4423/24/25 drivers can also be driven directly by the
SG1524/25/26/27, TL494/95, TL594/95, NE5560/61/62/68,
TSC170, MIC38C42, and similar switch mode power supply
ICs. By relocating the main switch drive function into the driver
rather than using the somewhat limited drive capabilities of a
PWM IC. The PWM IC runs cooler, which generally improves
its performance and longevity, and the main switches switch
faster, which reduces switching losses and increase system
efficiency.
The input protection circuitry of the MIC4423/24/25, in addi-
tion to providing 2kV or more of ESD protection, also works to
prevent latchup or logic upset due to ringing or voltage spiking
on the logic input terminal. In most CMOS devices when the
logic input rises above the power supply terminal, or descends
below the ground terminal, the device can be destroyed or
rendered inoperable until the power supply is cycled OFF
and ON. The MIC4423/24/25 drivers have been designed to
prevent this. Input voltages excursions as great as 5V below
ground will not alter the operation of the device. Input excur-
sions above the power supply voltage will result in the excess
voltage being conducted to the power supply terminal of the
IC. Because the excess voltage is simply conducted to the
power terminal, if the input to the driver is left in a high state
when the power supply to the driver is turned off, currents as
high as 30mA can be conducted through the driver from the
input terminal to its power supply terminal. This may overload
the output of whatever is driving the driver, and may cause
other devices that share the driver’s power supply, as well as
the driver, to operate when they are assumed to be off, but
it will not harm the driver itself. Excessive input voltage will
also slow the driver down, and result in much longer internal
propagation delays within the drivers. T
D2
, for example, may
increase to several hundred nanoseconds. In general, while
the driver will accept this sort of misuse without damage,
proper termination of the line feeding the driver so that line
spiking and ringing are minimized, will always result in faster
and more reliable operation of the device, leave less EMI to
be filtered elsewhere, be less stressful to other components
in the circuit, and leave less chance of unintended modes of
operation.
Power Dissipation
CMOS circuits usually permit the user to ignore power dis-
sipation. Logic families such as 4000 series and 74Cxxx have
outputs which can only source or sink a few milliamps of cur-
rent, and even shorting the output of the device to ground or
V
CC
may not damage the device. CMOS drivers, on the other
hand, are intended to source or sink several Amps of current.
This is necessary in order to drive large capacitive loads at
frequencies into the megahertz range. Package power dis-
sipation of driver ICs can easily be exceeded when driving
large loads at high frequencies. Care must therefore be paid
to device dissipation when operating in this domain.
The Supply Current vs Frequency and Supply Current vs
Load characteristic curves furnished with this data sheet
aid in estimating power dissipation in the driver. Operating
July 2005 9 MIC4423/4424/4425
MIC4423/4424/4425 Micrel, Inc.
frequency, power supply voltage, and load all affect power
dissipation.
Given the power dissipation in the device, and the thermal
resistance of the package, junction operating temperature
for any ambient is easy to calculate. For example, the ther-
mal resistance of the 8-pin plastic DIP package, from the
datasheet, is 150°C/W. In a 25°C ambient, then, using a
maximum junction temperature of 150°C, this package will
dissipate 960mW.
Accurate power dissipation numbers can be obtained by sum-
ming the three sources of power dissipation in the device:
• Load power dissipation (P
L
)
• Quiescent power dissipation (P
Q
)
• Transition power dissipation (P
T
)
Calculation of load power dissipation differs depending on
whether the load is capacitive, resistive or inductive.
Resistive Load Power Dissipation
Dissipation caused by a resistive load can be calculated as:
P
L
= I
2
R
O
D
where:
I = the current drawn by the load
R
O
= the output resistance of the driver when the
output is high, at the power supply voltage used
(See characteristic curves)
D = fraction of time the load is conducting (duty cycle)
Capacitive Load Power Dissipation
Dissipation caused by a capacitive load is simply the energy
placed in, or removed from, the load capacitance by the
driver. The energy stored in a capacitor is described by the
equation:
E = 1/2 C V
2
As this energy is lost in the driver each time the load is charged
or discharged, for power dissipation calculations the 1/2 is
removed. This equation also shows that it is good practice
not to place more voltage in the capacitor than is necessary,
as dissipation increases as the square of the voltage applied
to the capacitor. For a driver with a capacitive load:
P
L
= f C (V
S
)
2
where:
f = Operating Frequency
C = Load Capacitance
V
S
= Driver Supply Voltage
Inductive Load Power Dissipation
For inductive loads the situation is more complicated. For
the part of the cycle in which the driver is actively forcing
current into the inductor, the situation is the same as it is in
the resistive case:
P
L1
= I
2
R
O
D
However, in this instance the R
O
required may be either the on
resistance of the driver when its output is in the high state, or
its on resistance when the driver is in the low state, depending
on how the inductor is connected, and this is still only half the
story. For the part of the cycle when the inductor is forcing
current through the driver, dissipation is best described as
P
L2
= I V
D
(1 – D)
where V
D
is the forward drop of the clamp diode in the driver
(generally around 0.7V). The two parts of the load dissipation
must be summed in to produce P
L
P
L
= P
L1
+ P
L2
Quiescent Power Dissipation
Quiescent power dissipation (P
Q
, as described in the input
section) depends on whether the input is high or low. A low
input will result in a maximum current drain (per driver) of
≤0.2mA; a logic high will result in a current drain of ≤2.0mA.
Quiescent power can therefore be found from:
P
Q
= V
S
[D I
H
+ (1 – D) I
L
]
where:
I
H
= quiescent current with input high
I
L
= quiescent current with input low
D = fraction of time input is high (duty cycle)
V
S
= power supply voltage
Transition Power Dissipation
Transition power is dissipated in the driver each time its
output changes state, because during the transition, for a
very brief interval, both the N- and P-channel MOSFETs in
the output totem-pole are ON simultaneously, and a current
is conducted through them from V
S
to ground. The transition
power dissipation is approximately:
P
T
= f V
S
(A•s)
where (A•s) is a time-current factor derived from Figure 2.
Total power (PD) then, as previously described is just
P
D
= P
L
+ P
Q
+P
T
Examples show the relative magnitude for each term.
EXAMPLE 1: A MIC4423 operating on a 12V supply driving
two capacitive loads of 3000pF each, operating at 250kHz,
with a duty cycle of 50%, in a maximum ambient of 60°C.
First calculate load power loss:
P
L
= f x C x (V
S
)
2
P
L
= 250,000 x (3 x 10
–9
+ 3 x 10
–9
) x 12
2
= 0.2160W
Then transition power loss:
P
T
= f x V
S
x (A•s)
= 250,000 • 12 • 2.2 x 10
–9
= 6.6mW
P1-P3P4-P6P7-P9P10-P12P13-P13

MIC4424YM

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Manufacturer:
Microchip Technology / Micrel
Description:
Gate Drivers 3A Dual High Speed MOSFET Driver
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