OMO ElectronicOMO Electronic
Expand menu
Hello, Sign inMy Account
0 Cart

LTC1475IS8#PBF

P1-P3P4-P6P7-P9P10-P12P13-P15P16-P18P19-P20
7
LTC1474/LTC1475
APPLICATIONS INFORMATION
WUU
U
ments. Lower peak currents have the advantage of lower
output ripple (∆V
OUT
= I
PEAK
• ESR), lower noise, and less
stress on alkaline batteries and other circuit components.
Also, lower peak currents allow the use of inductors with
smaller physical size.
Peak currents as low as 10mA can be programmed with
the appropriate sense resistor. Increasing R
SENSE
above
10Ω, however, gives no further reduction of I
PEAK
.
For R
SENSE
values less than 1Ω, it is recommended that
the user parallel standard resistors (available in values ≥
1Ω) instead of using a special low valued shunt resistor.
Although a single resisor could be used with the desired
value, these low valued shunt resistor types are much
more expensive and are currently not available in case
sizes smaller than 1206. Three or four 0603 size standard
resistors require about the same area as one 1206 size
current shunt resistor at a fraction of the cost.
At higher supply voltages and lower inductances, the peak
currents may be slightly higher due to current comparator
overshoot and can be predicted from the second term in
the following equation:
I
R
VV
L
PEAK
SENSE
IN OUT
=
+
+
()
−
()
−
01
025
25 10
7
.
.
.
(2)
Note that R
SENSE
only sets the maximum inductor current
peak. At lower dI/dt (lower input voltages and higher
inductances), the observed peak current at loads less than
I
MAX
may be less than this calculated peak value due to the
voltage comparator tripping before the current ramps up
high enough to trip the current comparator. This effect
improves efficiency at lower loads by keeping the I
2
R
losses down (see Efficiency Considerations section).
Inductor Value Selection
Once R
SENSE
and I
PEAK
are known, the inductor value can
be determined. The minimum inductance recommended
as a function of I
PEAK
and I
MAX
can be calculated from:
L
VVt
II
MIN
OUT D OFF
PEAK MAX
≥
+
()
−
075.
(3)
where t
OFF
= 4.75µs.
MAXIMUM OUTPUT CURRENT (mA)
0
R
SENSE
(Ω)
5
4
3
2
1
0
50
100 150 200
1474/75 F02
250 300
FOR LOWEST NOISE
FOR BEST EFFICIENCY
Figure 2. R
SENSE
Selection
The basic LTC1474/LTC1475 application circuit is shown
in Figure 1, a high efficiency step-down converter. External
component selection is driven by the load requirement and
begins with the selection of R
SENSE
. Once R
SENSE
is
known, L can be chosen. Finally D1, C
IN
and C
OUT
are
selected.
R
SENSE
Selection
The current sense resistor (R
SENSE
) allows the user to
program the maximum inductor/switch current to opti-
mize the inductor size for the maximum load. The LTC1474/
LTC1475 current comparator has a maximum threshold of
100mV/(R
SENSE
+ 0.25). The maximum average output
current I
MAX
is equal to this peak value less half the peak-
to-peak ripple current ∆I
L
.
Allowing a margin for variations in the LTC1474/LTC1475
and external components, the required R
SENSE
can be
calculated from Figure 2 and the following equation:
R
SENSE
= (0.067/I
MAX
) – 0.25 (1)
for 10mA < I
MAX
< 200mA.
For I
MAX
above 200mA, R
SENSE
is set to zero by shorting
Pins 6 and 7 to provide the maximum peak current of
400mA (limited by the fixed internal sense resistor). This
400mA default peak current can be used for lower I
MAX
if
desired to eliminate the need for the sense resistor and
associated decoupling capacitor. However, for optimal
performance, the peak inductor current should be set to no
more than what is needed to meet the load current require-
8
LTC1474/LTC1475
APPLICATIONS INFORMATION
WUU
U
If the L
MIN
calculated is not practical, a larger I
PEAK
should
be used. Although the above equation provides the mini-
mum, better performance (efficiency, line/load regulation,
noise) is usually gained with higher values. At higher
inductances, peak current and frequency decrease (im-
proving efficiency) and inductor ripple current decreases
(improving noise and line/load regulation). For a given
inductor type, however, as inductance is increased, DC
resistance (DCR) increases, increasing copper losses,
and current rating decreases, both effects placing an
upper limit on the inductance. The recommended range of
inductances for small surface mount inductors as a func-
tion of peak current is shown in Figure 3. The values in this
range are a good compromise between the trade-offs
discussed above. If space is not a premium, inductors with
larger cores can be used, which extends the recom-
mended range of Figure 3 to larger values.
section, increased inductance requires more turns of wire
and therefore copper losses will increase.
Ferrite and Kool Mµ
designs have very low core loss and
are preferred at high switching frequencies, so design
goals can concentrate on copper loss and preventing
saturation. Ferrite core material saturates “hard,” which
means that inductance collapses abruptly when the peak
design current is exceeded. This results in an abrupt
increase in inductor current above I
PEAK
and consequent
increase in voltage ripple.
Do not allow the core to satu-
rate!
Coiltronics, Coilcraft, Dale and Sumida make high
performance inductors in small surface mount packages
with low loss ferrite and Kool Mµ cores and work well in
LTC1474/LTC1475 regulators.
Catch Diode Selection
The catch diode carries load current during the off-time.
The average diode current is therefore dependent on the
P-channel switch duty cycle. At high input voltages the
diode conducts most of the time. As V
IN
approaches V
OUT
the diode conducts only a small fraction of the time. The
most stressful condition for the diode is when the output
is short-circuited. Under this condition, the diode must
safely handle I
PEAK
at close to 100% duty cycle.
To maximize both low and high current efficiency, a fast
switching diode with low forward drop and low reverse
leakage should be used. Low reverse leakage current is
critical to maximize low current efficiency since the leak-
age can potentially approach the magnitude of the LTC1474/
LTC1475 supply current. Low forward drop is critical for
high current efficiency since loss is proportional to for-
ward drop. These are conflicting parameters (see Table 1),
but a good compromise is the MBR0530 0.5A Schottky
diode specified in the application circuits.
Table 1. Effect of Catch Diode on Performance
FORWARD NO LOAD
DIODE (D1) LEAKAGE DROP SUPPLY CURRENT EFFICIENCY*
BAS85 200nA 0.6V 9.7µA 77.9%
MBR0530 1µA 0.4V 10µA 83.3%
MBRS130 20µA 0.3V 16µA 84.6%
*Figure 1 circuit with V
IN
= 15V, I
OUT
= 0.1A
Kool Mµ is a registered trademark of Magnetics, Inc.
PEAK INDUCTOR CURRENT (mA)
10
50
500
INDUCTOR VALUE (µH)
100
1000
100 1000
1474/75 F03
Inductor Core Selection
Once the value of L is known, the type of inductor must be
selected. High efficiency converters generally cannot
afford the core loss found in low cost powdered iron cores,
forcing the use of more expensive ferrite, molypermalloy
or Kool Mµ
®
cores. Actual core loss is independent of core
size for a fixed inductor value, but is very dependent on
inductance selected. As inductance increases, core losses
go down. Unfortunately, as discussed in the previous
Figure 3. Recommended Inductor Values
9
LTC1474/LTC1475
APPLICATIONS INFORMATION
WUU
U
C
IN
and C
OUT
Selection
At higher load currents, when the inductor current is
continuous, the source current of the P-channel MOSFET
is a square wave of duty cycle V
OUT
/V
IN
. To prevent large
voltage transients, a low ESR input capacitor sized for the
maximum RMS current must be used. The maximum
capacitor current is given by:
C
VVV
V
IN
OUT IN OUT
IN
Required I =
I
RMS
MAX
−
()
[]
12/
This formula has a maximum at V
IN
= 2V
OUT
, where
I
RMS
= I
OUT
/2. This simple worst-case condition is com-
monly used for design because even significant deviations
do not offer much relief. Note that capacitor manufacturer’s
ripple current ratings are often based on 2000 hours of life.
This makes it advisable to further derate the capacitor, or
to choose a capacitor rated at a higher temperature than
required. Do not underspecify this component. An addi-
tional 0.1µF ceramic capacitor is also required on V
IN
for
high frequency decoupling.
The selection of C
OUT
is driven by the required effective
series resistance (ESR) to meet the output voltage ripple
and line regulation requirements. The output voltage ripple
during a burst cycle is dominated by the output capacitor
ESR and can be estimated from the following relation:
25mV < ∆V
OUT, RIPPLE
= ∆I
L
• ESR
where ∆I
L
≤ I
PEAK
and the lower limit of 25mV is due to the
voltage comparator hysteresis. Line regulation can also
vary with C
OUT
ESR in applications with a large input
voltage range and high peak currents.
ESR is a direct function of the volume of the capacitor.
Manufacturers such as Nichicon, AVX and Sprague should
be considered for high performance capacitors. The
OS-CON semiconductor dielectric capacitor available from
SANYO has the lowest ESR for its size at a somewhat
higher price. Typically, once the ESR requirement is satis-
fied, the capacitance is adequate for filtering. For lower
current applications with peak currents less than 50mA,
10µF ceramic capacitors provide adequate filtering and
are a good choice due to their small size and almost
negligible ESR. AVX and Marcon are good sources for
these capacitors.
In surface mount applications multiple capacitors may
have to be paralleled to meet the ESR or RMS current
handling requirements of the application. Aluminum elec-
trolytic and dry tantalum capacitors are both available in
surface mount configurations. In the case of tantalum, it is
critical that the capacitors are surge tested for use in
switching power supplies. An excellent choice is the AVX
TPS series of surface mount tantalums, available in case
heights ranging from 2mm to 4mm. Other capacitor types
include SANYO OS-CON, Nichicon PL series and Sprague
595D series. Consult the manufacturer for other specific
recommendations.
To avoid overheating, the output capacitor must be sized
to handle the ripple current generated by the inductor. The
worst-case ripple current in the output capacitor is given
by:
I
RMS
= I
PEAK
/2
Once the ESR requirement for C
OUT
has been met, the
RMS current rating generally far exceeds the I
RIPPLE(P-P)
requirement.
Efficiency Considerations
The efficiency of a switching regulator is equal to the
output power divided by the input power times 100%. It is
often useful to analyze individual losses to determine what
is limiting efficiency and which change would produce the
most improvement. Efficiency can be expressed as:
Efficiency = 100% – (L1 + L2 + L3 + ...)
where L1, L2, etc. are the individual losses as a percentage
of input power.
Although all dissipative elements in the circuit produce
losses, three main sources usually account for most of the
losses in LTC1474/LTC1475 circuits: V
IN
current, I
2
R
losses and catch diode losses.
1. The V
IN
current is due to two components: the DC bias
current and the internal P-channel switch gate charge
current. The DC bias current is 9µA at no load and
increases proportionally with load up to a constant
100µA during continuous mode. This bias current is so
P1-P3P4-P6P7-P9P10-P12P13-P15P16-P18P19-P20

LTC1475IS8#PBF

Mfr. #:
Buy LTC1475IS8#PBF
Manufacturer:
Analog Devices / Linear Technology
Description:
Switching Voltage Regulators Low IQ Push-Button Stepdn DC/DC Conv.
Lifecycle:
New from this manufacturer.
Delivery:
DHL FedEx Ups TNT EMS
Payment:
T/T Paypal Visa MoneyGram Western Union

Products related to this Datasheet