AD629
Rev. C | Page 12 of 16
ANALOG POWER
SUPPLY
DIGITAL
POWER SUPPLY
0.1µF
0.1µF
0.1µF0.1µF
+IN
–IN
–V
S
V
IN1
V
IN2
V
DD
V
DD
OUTPUT
AGND
GND
MICROPROCESSOR
DGND
+V
S
AD629
AD7892-2
REF(–) REF(+)
6
7
14
4
1
3
3
2
6 4
1 5
12
+5V GND
+5VGND
–5V
00783-032
Figure 34. Optimal Grounding Practice for a Bipolar Supply Environment
with Separate Analog and Digital Supplies
POWER SUPPLY
V
IN1
V
IN2
V
DD
AGND DGND
ADC
0.1µF
0.1µF
+IN
–IN
+V
S
OUTPUT
–V
S
AD629
REF(–) REF(+)
47
3
2
6
1 5
V
DD
GND
MICROPROCESSOR
+5V
GND
0.1µF
00783-033
Figure 35. Optimal Ground Practice in a Single-Supply Environment
If there is only a single power supply available, it must be shared
by both digital and analog circuitry. Figure 35 shows how to
minimize interference between the digital and analog circuitry.
In this example, the ADC’s reference is used to drive Pin REF(+)
and Pin REF(–). This means that the reference must be capable
of sourcing and sinking a current equal to V
CM
/200 k. As in
the previous case, separate analog and digital ground planes
should be used (reasonably thick traces can be used as an
alternative to a digital ground plane). These ground planes
should connect at the power supply’s ground pin. Separate
traces (or power planes) should run from the power supply to
the supply pins of the digital and analog circuits. Ideally, each
device should have its own power supply trace, but these can be
shared by a number of devices, as long as a single trace is not
used to route current to both digital and analog circuitry.
USING A LARGE SENSE RESISTOR
Insertion of a large value shunt resistance across the input pins,
Pin 2 and Pin 3, will imbalance the input resistor network,
introducing a common-mode error. The magnitude of the error
will depend on the common-mode voltage and the magnitude
of R
SHUNT
.
Tabl e 5 shows some sample error voltages generated by a
common-mode voltage of 200 V dc with shunt resistors from
20 to 2000 . Assuming that the shunt resistor is selected to
use the full ±10 V output swing of the AD629, the error voltage
becomes quite significant as R
SHUNT
increases.
Table 5. Error Resulting from Large Values of R
SHUNT
(Uncompensated Circuit)
R
S
(Ω) Error V
OUT
(V) Error Indicated (mA)
20 0.01 0.5
1000 0.498 0.498
2000 1 0.5
To measure low current or current near zero in a high common-
mode environment, an external resistor equal to the shunt
resistor value can be added to the low impedance side of the
shunt resistor, as shown in Figure 36.
REF (–)
REF (+)
–V
S
–V
S
+V
S
+V
S
V
OUT
NC
–IN
+IN
R
SHUNT
R
COMP
I
SHUNT
0.1µF
0.1µF
NC = NO CONNECT
21.1kΩ
380kΩ 380kΩ
20kΩ
380kΩ
AD629
1
2
3
4
8
7
6
5
00783-034
Figure 36. Compensating for Large Sense Resistors
OUTPUT FILTERING
A simple 2-pole, low-pass Butterworth filter can be implemented
using the OP177 after the AD629 to limit noise at the output, as
shown in Figure 37. Table 6 gives recommended component
values for various corner frequencies, along with the peak-to-
peak output noise for each case.
REF (–)
REF (+)
V
S
–V
S
+V
S
+V
S
+V
S
V
OUT
NC
–IN
+IN
0.1µF
0.1µF 0.1µF
0.1µF
NC = NO CONNECT
21.1kΩ
380kΩ 380kΩ
20kΩ
380kΩ
AD629
1
2
3
4
8
7
6
5
00783-035
R1 R2
C1
C2
OP177
Figure 37. Filtering of Output Noise Using a 2-Pole Butterworth Filter
Table 6. Recommended Values for 2-Pole Butterworth Filter
Corner Frequency R1 R2 C1 C2 Output Noise (p-p)
No Filter
3.2 mV
50 kHz 2.94 kΩ ± 1% 1.58 kΩ ± 1% 2.2 nF ± 10% 1 nF ± 10% 1 mV
5 kHz 2.94 kΩ ± 1% 1.58 kΩ ± 1% 22 nF ± 10% 10 nF ± 10% 0.32 mV
500 Hz 2.94 kΩ ± 1% 1.58 kΩ ± 1% 220 nF ± 10% 0.1 μF ± 10% 100 μV
50 Hz 2.7 kΩ ± 10% 1.5 kΩ ± 10% 2.2 μF ± 20% 1 μF ± 20% 32 μV
