MIC2198YML Datasheet MIC2198YML, MIC2198YML-TR, MIC2198BML-TR | P7-P9

October 2005 7 MIC2198

MIC2198 Micrel, Inc.

Current Limit

The MIC2198 output current is detected by the voltage drop

across the external current-sense resistor (R

in Figure

2.). The current limit threshold is 75mV±25mV. The current-

sense resistor must be sized using the minimum current

limit threshold. The external components must be designed

to withstand the maximum current limit. The current-sense

resistor value is calculated by the equation below:

55mV

OUT(max)

The maximum output current is:

95mV

OUT(max)

The current-sense pins CSH (pin 4) and V

OUT

(pin 5) are

noise sensitive due to the low signal level and high input im-

pedance. The PCB traces should be short and routed close

to each other. A small (1nF to 0.1µF) capacitor across the

pins will attenuate high frequency switching noise.

When the peak inductor current exceeds the current limit

threshold, the current limit comparator, in Figure 2, turns off

the high-side MOSFET for the remainder of the cycle. The

output voltage drops as additional load current is pulled from

the converter. When the output voltage reaches approximately

0.4V, the circuit enters frequency-foldback mode and the

oscillator frequency will drop to 125kHz while maintaining the

peak inductor current equal to the nominal 75mV across the

external current-sense resistor. This limits the maximum output

power delivered to the load under a short circuit condition.

Reference, Enable and UVLO Circuits

The output drivers are enabled when the following conditions

are satisﬁed:

• The V

voltage (pin 7) is greater than its under-

voltage threshold (typically 4.25V).

• The voltage on the enable pin is greater than the

enable UVLO threshold (typically 2.5V).

The internal bias circuit generates a 0.8V bandgap reference

voltage for the voltage error ampliﬁer and a 5V V

voltage

for the gate drive circuit. The MIC2198 uses FB (pin 3) for

output voltage sensing.

The enable pin (pin 2) has two threshold levels, allowing

the MIC2198 to shut down in a low current mode, or turn off

output switching in UVLO mode. An enable pin voltage lower

than the shutdown threshold turns off all the internal circuitry

and reduces the input current to typically 0.1µA.

If the enable pin voltage is between the shutdown and UVLO

thresholds, the internal bias, V

, and reference voltages are

turned on. The output drivers are inhibited from switching and

remain in a low state. Raising the enable voltage above the

UVLO threshold of 2.5V enables the output drivers.

Either of two UVLO conditions will disable the MIC2198 from

switching.

• When the V

drops below 4.1V

• When the enable pin drops below the 2.5V threshold

MOSFET Gate Drive

The MIC2198 high-side drive circuit is designed to switch

an N-Channel MOSFET. Referring to the block diagram in

Figure 2, a bootstrap circuit, consisting of D2 and C

BST

, sup-

plies energy to the high-side drive circuit. Capacitor C

BST

charged while the low-side MOSFET is on and the voltage on

the V

pin (pin 11) is approximately 0V. When the high-side

MOSFET driver is turned on, energy from C

BST

is used to

turn the MOSFET on. As the MOSFET turns on, the voltage

on the V

pin increases to approximately V

. Diode D2

is reversed biased and C

BST

ﬂoats high while continuing to

keep the high-side MOSFET on. When the low-side switch

is turned back on, C

BST

is recharged through D2.

The drive voltage is derived from the internal 5V V

bias

supply. The nominal low-side gate drive voltage is 5V and

the nominal high-side gate drive voltage is approximately

4.5V due the voltage drop across D2. A ﬁxed 80ns delay

between the high- and low-side driver transitions is used

to prevent current from simultaneously ﬂowing unimpeded

through both MOSFETs.

Oscillator

The internal oscillator is free running and requires no external

components. The nominal oscillator frequency is 500kHz. If

the output voltage is below approximately 0.4V, the oscillator

operates in a frequency-foldback mode and the switching

frequency is reduced to 125kHz.

OUT

TIME

= 7V

= 125kHz

OUT

= 0.4V

= 500kHz

OUT

= 3.3V

Figure 4. Startup Waveform

Above 0.4V, the switching frequency increases to 500kHz

causing the output voltage to rise a greater rate. The rise

time of the output is dependent on the output capacitance,

output voltage, and load current. The oscilloscope photo in

Figure 4 show the output voltage at startup.

MIC2198 Micrel, Inc.

MIC2198 8 October 2005

Minimum Pulsewidth

The MIC2198 has a speciﬁed minimum pulsewidth. This

minimum pulsewidth places a lower limit on the minimum

duty cycle of the buck converter.

Figure 5 shows the minimum output voltage versus input

supply voltage for the MIC2198. For example, for V

= 15V,

OUT

= 1.65V would be the lowest achievable voltage that

conforms to the minimum-on-time.

0.0

0.5

1.0

1.5

2.0

2.5

3.0

3.5

4.0

4.5 9.5 14.5 19.5 24.5 29.5

INPUT VOLTAGE (V)

Figure 5. Minimum Output Voltage

vs. Input Supply Voltage

October 2005 9 MIC2198

MIC2198 Micrel, Inc.

Applications Information

The following applications information includes component

selection and design guidelines.

Inductor Selection

Values for inductance, peak, and RMS currents are required

to select the output inductor. The input and output voltages

and the inductance value determine the peak to peak induc-

tor ripple current. Generally, higher inductance values are

used with higher input voltages. Larger peak to peak ripple

currents will increase the power dissipation in the inductor

and MOSFETs. Larger output ripple currents will also require

more output capacitance to smooth out the larger ripple cur-

rent. Smaller peak to peak ripple currents require a larger

inductance value and therefore a larger and more expensive

inductor. A good compromise between size, loss and cost is

to set the inductor ripple current to be equal to 20% of the

maximum output current.

The inductance value is calculated by the equation below.

OUT

× (V

IN(max)

− V

OUT

)

IN(max)

× f

× 0.2 × I

OUT(max)

where:

= switching frequency

0.2 = ratio of AC ripple current to DC output current

IN(max)

= maximum input voltage

The peak-to-peak inductor current (AC ripple current) is:

OUT

× (V

IN(max)

−V

OUT

)

IN(max)

× f

× L

The peak inductor current is equal to the average output current

plus one half of the peak to peak inductor ripple current.

I I 0.5 I

PK OUT(max) PP

= + ×

The RMS inductor current is used to calculate the I

×R losses

in the inductor.

I I 1

INDUCTOR(rms) OUT(max)

OUT(max)

= × +













Maximizing efﬁciency requires the proper selection of core

material and minimizing the winding resistance. The high

frequency operation of the MIC2198 requires the use of fer-

rite materials for all but the most cost sensitive applications.

Lower cost iron powder cores may be used but the increase

in core loss will reduce the efﬁciency of the power supply.

This is especially noticeable at low output power. The winding

resistance decreases efﬁciency at the higher output current

levels. The winding resistance must be minimized although

this usually comes at the expense of a larger inductor.

The power dissipated in the inductor is equal to the sum

of the core and copper losses. At higher output loads, the

core losses are usually insigniﬁcant and can be ignored. At

lower output currents, the core losses can be a signiﬁcant

contributor. Core loss information is usually available from

the magnetics vendor.

Copper loss in the inductor is calculated by the equation

below:

INDUCTORCu

= I

INDUCTOR(rms)

× R

WINDING

The resistance of the copper wire, R

WINDING

, increases with

temperature. The value of the winding resistance used should

be at the operating temperature.

WINDING(hot)

= R

WINDING(20°C)

× 1 + 0.0042 × (T

HOT

− T

20°C

)













where:

HOT

= temperature of the wire under operating load

20°C

= ambient temperature

WINDING(20°C)

is room temperature winding

resistance (usually speciﬁed by the manufacturer)

Current-Sense Resistor Selection

Low inductance power resistors, such as metal ﬁlm resistors

should be used. Most resistor manufacturers make low induc-

tance resistors with low temperature coefﬁcients, designed

speciﬁcally for current-sense applications. Both resistance

and power dissipation must be calculated before the resis-

tor is selected. The value of R

SENSE

is chosen based on the

maximum output current and the maximum threshold level.

The power dissipated is based on the maximum peak output

current at the minimum overcurrent threshold limit.

55mV

SENSE

OUT(max)

The maximum overcurrent threshold is:

95mV

OVERCURRENT(max)

The maximum power dissipated in the sense resistor is:

D(R

SENCE

)

= I

OVERCURRENT(max)

× R

MOSFET Selection

External N-Channel logic-level power MOSFETs must be

used for the high- and low-side switches. The MOSFET

gate-to-source drive voltage of the MIC2198 is regulated by

an internal 5V V

regulator. Logic-level MOSFETs, whose

operation is speciﬁed at V

= 4.5V must be used.

It is important to note the on-resistance of a MOSFET in-

creases with increasing temperature. A 75°C rise in junction

temperature will increase the channel resistance of the MOS-

FET by 50% to 75% of the resistance speciﬁed at 25°C. This

change in resistance must be accounted for when calculating

MOSFET power dissipation.

Total gate charge is the charge required to turn the MOSFET

on and off under speciﬁed operating conditions (V

and

). The gate charge is supplied by the MIC2198 gate drive

circuit. At 500kHz switching frequency, the gate charge can

be a signiﬁcant source of power dissipation in the MIC2198.

At low output load this power dissipation is noticeable as a

reduction in efﬁciency. The average current required to drive

the high-side MOSFET is:

I Q f

G[high-side](avg) G S

= ×

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MIC2198YML

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Switching Controllers 500KHz Sync Buck Controller

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