5
FN3395.8
June 6, 2006
Application Information
Optimum Feedback Resistor
Although a current feedback amplifier’s bandwidth
dependency on closed loop gain isn’t as severe as that of a
voltage feedback amplifier, there can be an appreciable
decrease in bandwidth at higher gains. This decrease may
be minimized by taking advantage of the current feedback
amplifier’s unique relationship between bandwidth and R
F
.
All current feedback amplifiers require a feedback resistor,
even for unity gain applications, and R
F
, in conjunction with
the internal compensation capacitor, sets the dominant pole
of the frequency response. Thus, the amplifier’s bandwidth is
inversely proportional to R
F
. The HFA1105 design is
optimized for R
F
= 510Ω at a gain of +2. Decreasing R
F
decreases stability, resulting in excessive peaking and
overshoot (Note: Capacitive feedback will cause the same
problems due to the feedback impedance decrease at higher
frequencies). At higher gains, however, the amplifier is more
stable so R
F
can be decreased in a trade-off of stability for
bandwidth.
The table below lists recommended R
F
values for various
gains, and the expected bandwidth. For a gain of +1, a
resistor (
+R
S
) in series with +IN is required to reduce gain
peaking and increase stability.
Non-Inverting Input Source Impedance
For best operation, the DC source impedance seen by the
non-inverting input should be ≥50Ω. This is especially
important in inverting gain configurations where the non-
inverting input would normally be connected directly to GND.
Pulse Undershoot and Asymmetrical Slew Rates
The HFA1105 utilizes a quasi-complementary output stage to
achieve high output current while minimizing quiescent supply
current. In this approach, a composite device replaces the
traditional PNP pulldown transistor. The composite device
switches modes after crossing 0V, resulting in added
distortion for signals swinging below ground, and an
increased undershoot on the negative portion of the output
waveform (See Figures 5, 8, and 11). This undershoot isn’t
present for small bipolar signals, or large positive signals.
Another artifact of the composite device is asymmetrical slew
rates for output signals with a negative voltage component.
The slew rate degrades as the output signal crosses through
0V (See Figures 5, 8, and 11), resulting in a slower overall
negative slew rate. Positive only signals have symmetrical
slew rates as illustrated in the large signal positive pulse
response graphs (See Figures 4, 7, and 10).
PC Board Layout
The amplifier’s frequency response depends greatly on the
care taken in designing the PC board. The use of low
inductance components such as chip resistors and chip
capacitors is strongly recommended, while a solid
ground plane is a must!
Attention should be given to decoupling the power supplies.
A large value (10µF) tantalum in parallel with a small value
(0.1µF) chip capacitor works well in most cases.
Terminated microstrip signal lines are recommended at the
device’s input and output connections. Capacitance,
parasitic or planned, connected to the output must be
minimized, or isolated as discussed in the next section.
Care must also be taken to minimize the capacitance to
ground at the amplifier’s inverting input (-IN), as this
capacitance causes gain peaking, pulse overshoot, and if
large enough, instability. To reduce this capacitance, the
designer should remove the ground plane under traces
connected to
-IN, and keep connections to -IN as short as possible.
An example of a good high frequency layout is the
Evaluation Board shown in Figure 2.
Driving Capacitive Loads
Capacitive loads, such as an A/D input, or an improperly
terminated transmission line will degrade the amplifier’s
phase margin resulting in frequency response peaking and
possible oscillations. In most cases, the oscillation can be
avoided by placing a resistor (R
S
) in series with the output
prior to the capacitance.
Figure 1 details starting points for the selection of this
resistor. The points on the curve indicate the R
S
and C
L
combinations for the optimum bandwidth, stability, and
settling time, but experimental fine tuning is recommended.
Picking a point above or to the right of the curve yields an
overdamped response, while points below or left of the curve
indicate areas of underdamped performance.
R
S
and C
L
form a low pass network at the output, thus limiting
system bandwidth well below the amplifier bandwidth of
270MHz (for A
V
= +1). By decreasing R
S
as C
L
increases (as
illustrated in the curves), the maximum bandwidth is obtained
without sacrificing stability. In spite of this, the bandwidth
decreases as the load capacitance increases. For example, at
A
V
= +1, R
S
= 62Ω, C
L
= 40pF, the overall bandwidth is limited
to 180MHz, and bandwidth drops to 75MHz at A
V
= +1,
R
S
=8Ω, C
L
= 400pF.
GAIN
(A
CL
)R
F
(Ω)
BANDWIDTH
(MHz)
-1 425 300
+1 510 (+R
S
= 510Ω) 270
+2 510 330
+5 200 300
+10 180 130
HFA1105
