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LTC3873EDDB#TRPBF

P1-P3P4-P6P7-P9P10-P12P13-P15P16-P18
LTC3873
10
3873fb
For more information www.linear.com/LTC3873
The circuit in Figure 6 shows a third way to power the
LTC3873. An external series pre-regulator consisting of
series pass transistor Q1, zener diode D1 and bias resis
-
tor R
B
brings V
CC
to at least 7.6V nominal, well above
the maximum rated V
CC
turn-off threshold of 4V. Resistor
R
START
momentarily charges the V
CC
node up to the V
CC
turn-on threshold, enabling the LTC3873.
APPLICATIONS INFORMATION
ringing on the SW pin disrupts the tiny slope compensa-
tion current out of the pin. It is not recommended to add
external slope compensation in this case.
Output Voltage Programming
The output voltage is set by a resistor divider according
to the following formula:
V
O
= 1.2V • 1+
R2
R1
⎛
⎝
⎜
⎞
⎠
⎟
The external resistor divider is connected to the output
as shown in Figure 5, allowing remote voltage sensing.
Choose resistance values for R1 and R2 to be as large as
possible in order to minimize any efficiency loss due to
the static current drawn from V
OUT
, but just small enough
so that when V
OUT
is in regulation, the error caused by
the nonzero input current to the V
FB
pin is less than 1%.
A good rule of thumb is to choose R1 to be 24k or less.
Transformer Design Considerations
Transformer specification and design is perhaps the
most critical part of applying the LTC3873 successfully.
In addition to the usual list of caveats dealing with high
frequency power transformer design, the following should
prove useful.
Turns Ratios
Due to the use of the external feedback resistor divider
ratio to set output voltage, the user has relative freedom
in selecting a transformer turns ratio to suit a given ap-
plication. Simple ratios of small integers, e.g., 1:1, 2:1, 3:2,
etc. can be employed which yield more freedom in setting
total turns and mutual inductance. Simple integer turns
ratios also facilitate the use of “off-the-shelf” configurable
transformers such as the Coiltronics VERSA-PAC series
in applications with high input-to-output voltage ratios.
For example, if a 6-winding VERSA-PAC is used with three
windings in series on the primary and three windings in
parallel on the secondary, a 3:1 turns ratio will be achieved.
Turns ratio can be chosen on the basis of desired duty
cycle. However, remember that the input supply voltage
LTC3873
V
CC
3873 F06
Q1
R
B
R
START
V
IN
C
VCC
0.1µF
D1
8.2V
GND
Figure 6
Slope Compensation
The LTC3873 has built-in internal slope compensation to
stabilize the control loop against sub-harmonic oscillation.
It also provides the ability to externally increase slope
compensation by injecting a ramping current out of its SW
pin into an external slope compensation resistor (R
SL
in
Figure 5). This current ramp starts at zero right after the
NGATE pin has been set high. The current rises linearly
towards a peak of 20µA at the maximum duty cycle of
80%, shutting off once the NGATE pin goes low. A series
resistor (R
SL
) connecting the SW pin to the current sense
resistor (R
SENSE
) thus develops a ramping voltage drop.
From the perspective of the SW pin, this ramping voltage
adds to the voltage across the sense resistor, effectively
reducing the current comparator threshold in proportion
to duty cycle. The amount of reduction in the current
comparator threshold (ΔV
SENSE
) can be calculated using
the following equation:
ΔV
SENSE
=
DutyCycle – 6%
80%
20µA •R
SLOPE
Note the external programmable slope compensation is
only needed when the internal slope compensation is not
sufficient. In most applications R
SL
can be shorted. For the
LTC3873, when the R
DS(ON)
sensing technique is used, the
LTC3873
11
3873fb
For more information www.linear.com/LTC3873
plus the secondary-to-primary referred voltage of the
flyback pulse (including leakage spike) must not exceed
the allowed external MOSFET breakdown rating.
Leakage Inductance
Transformer leakage inductance (on either the primary
or secondary) causes a voltage spike to occur after the
output switch (Q1) turn-off. This is increasingly prominent
at higher load currents where more stored energy must
be dissipated. In some cases a “snubber” circuit will be
required to avoid overvoltage breakdown at the MOSFET’s
drain node. Application Note 19 is a good reference on
snubber design. A bifilar or similar winding technique is a
good way to minimize troublesome leakage inductances.
However, remember that this will limit the primary-to-
secondary breakdown voltage, so bifilar winding is not
always practical.
Power MOSFET Selection
The power MOSFET serves two purposes in the LTC3873:
it represents the main switching element in the power path
and its R
DS(ON)
represents the current sensing element
for the control loop. Important parameters for the power
MOSFET include the drain-to-source breakdown voltage
(BV
DSS
), the threshold voltage (V
GS(TH)
), the on-resistance
(R
DS(ON)
) versus gate-to-source voltage, the gate-to-source
and gate-to-drain charges (Q
GS
and Q
GD
, respectively),
the maximum drain current (I
D(MAX)
) and the MOSFET’s
thermal resistances (R
TH(JC)
and R
TH(JA)
).
For boost applications with R
DS(ON)
sensing, refer to
the LTC3872 data sheet for the selection of MOSFET
R
DS(ON)
.
MOSFETs have conduction losses (I
2
R) and switching
losses. For V
DS
< 20V, high current efficiency generally
improves with large MOSFETs with low R
DS(ON)
, while
for V
DS
> 20V the transition losses rapidly increase to the
point that the use of a higher R
DS(ON)
device with lower
reverse transfer capacitance, C
RSS
, actually provides
higher efficiency.
Output Capacitors
The output capacitor is normally chosen by its effective
series resistance (ESR), which determines output ripple
voltage and affects efficiency. Low ESR ceramic capaci
-
tors are often used to minimize the output ripple. Boost
regulators have large RMS ripple current in the output
capacitor that must be rated to handle the current. The
output ripple current (RMS) is:
I
RMS(COUT)
≈ I
OUT(MAX )
•
V
OUT
– V
IN(MIN)
V
IN(MIN)
Output ripple is then simply:
V
OUT
= R
ESR
(ΔI
L(RMS)
)
The output capacitor for flyback converter should have a
ripple current rating greater than:
I
RMS
= I
OUT
•
D
MAX
1– D
MAX
Input Capacitors
The input capacitor of a boost converter is less critical due
to the fact that the input current waveform is triangular, and
does not contain large square wave currents as found in
the output capacitor. The input voltage source impedance
determines the size of the capacitor that is typically 10μF
to 100μF. A low ESR is recommended although not as
critical as the output capacitor can be on the order of 0.3Ω.
The RMS input ripple current for a boost converter is:
I
RMS(CIN)
= 0.3 •
V
IN(MIN)
L • f
•D
MAX
Please note that the input capacitor can see a very high
surge current when a battery is suddenly connected to the
input of the converter and solid tantalum capacitors can
fail catastrophically under these conditions.
APPLICATIONS INFORMATION
LTC3873
12
3873fb
For more information www.linear.com/LTC3873
APPLICATIONS INFORMATION
In a flyback converter, the input flows in pulses placing
severe demands on the input capacitors. Select an input
capacitor with a ripple current rating greater than:
I
RMS
=
P
IN
V
IN(MIN)
1– D
MAX
D
MAX
Duty Cycle Considerations
The LTC3873 imposes a maximum duty cycle limit of
80% typical. For a flyback converter, the maximum duty
cycle prevents the transformer core from saturation. In
a boost converter application, however, it sets a limit on
the maximum step-up ratio or maximum output voltage
with the given input voltage of:
V
OUT(MAX )
=
V
IN(MIN)
1– 0.8
– V
D
Current and voltage stress on the power switch and
synchronous rectifiers, input and output capacitor RMS
currents and transformer utilization (size vs power) are
impacted by duty factor. Unfortunately duty factor can
-
not be adjusted to simultaneously optimize all of these
requirements. In general,
avoid extreme duty factors since
this severely impacts the current stress on most of the
components. A reasonable target for duty factor is 50% at
nominal input voltage. Using this rule of thumb, the ideal
transformer turns ratio is:
N
IDEAL
=
V
OUT
V
IN
•
1– D
D
=
V
OUT
V
IN
Output Diode Selection
To maximize efficiency, a fast switching diode with low
forward drop and low reverse leakage is desired. The output
diode in a boost converter conducts current during the
switch off-time. The peak reverse voltage that the diode
must withstand is equal to the regulator output voltage.
The average forward current in normal operation is equal
to the output current, and the peak current is equal to the
peak inductor current.
I
TH
NGATE
LTC3873
GND
RUN/SS
V
FB
= 1.2V
100µF
6.3V
×3
0.1µF
Q1
FAN2512
V
OUT
*
3.3V
3A
V
IN
36V TO 72V
4.7µF
100V
221k
4.7µF
10V
68mΩ
*FOR 5V OUTPUT CHANGE R
FB
TO 42.2k
15k
2.2nF
0.1µF
T1
D2
UPS840
D1
BAS516
3873 F07
•
•
V
CC
SW
IPRG
12.06k
V
OUT
R
FB
*
21.5k
51Ω
•
Figure 7. 3.3V Output Nonisolated Telecom DC/DC Converter
P1-P3P4-P6P7-P9P10-P12P13-P15P16-P18

LTC3873EDDB#TRPBF

Mfr. #:
Buy LTC3873EDDB#TRPBF
Manufacturer:
Analog Devices / Linear Technology
Description:
Switching Voltage Regulators No Rsense Constant Frequency Current Mode Boost/Flyback/SEPIC DC/DC Controller
Lifecycle:
New from this manufacturer.
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