Figure 9. Low Power Instrumentation Amplifier
Gains of 1 to 100 can be accommodated with gain nonlinearities
of less than 0.01%. Input errors, which contribute an output
error proportional to in amp gain, include a maximum untrimmed
input offset voltage of 0.5 mV and an input offset voltage drift
over temperature of 4 µV/°C. Output errors, which are indepen-
dent of gain, will contribute an additional 0.5 mV offset and
4 µV/°C drift. The maximum input current is 15 pA over the
common-mode range, with a common-mode impedance of over
1 × 10
12
Ω. Resistor pairs R3/R5 and R4/R6 should be ratio
matched to 0.01% to take full advantage of the AD548’s high
common-mode rejection. Capacitors C1 and C1′ compensate for
peaking in the gain over frequency caused by input capacitance
when gains of 1 to 3 are used.
The –3 dB small signal bandwidth for this low power instrumenta-
tion amplifier is 700 kHz for a gain of 1 and 10 kHz for a gain of
100. The typical output slew rate is 1.8 V/µs.
LOG RATIO AMPLIFIER
Log ratio amplifiers are useful for a variety of signal conditioning
applications, such as linearizing exponential transducer outputs
and compressing analog signals having a wide dynamic range.
The AD548’s picoamp level input current and low input offset
voltage make it a good choice for the front-end amplifier of the
log ratio circuit shown in Figure 10. This circuit produces an
output voltage equal to the log base 10 of the ratio of the input
currents I
1
and I
2
. Resistive inputs R1 and R2 are provided for
voltage inputs.
Input currents I
1
and I
2
set the collector currents of Q1 and Q2,
a matched pair of logging transistors. Voltages at points A and
B are developed according to the following familiar diode
equation:
V
BE
= (kT/q)ln(I
C
/I
ES
)
In this equation, k is Boltzmann’s constant, T is absolute tem-
perature, q is an electron charge, and I
ES
is the reverse saturation
current of the logging transistors. The difference of these two
voltages is taken by the subtractor section and scaled by a factor
of approximately 16 by resistors R9, R10, and R8. Temperature
Application Hints–AD548
PHOTODIODE PREAMP
The performance of the photodiode preamp shown in Figure 7
is enhanced by the AD548’s low input current, input voltage
offset, and offset voltage drift. The photodiode sources a current
proportional to the incident light power on its surface. R
F
converts
the photodiode current to an output voltage equal to R
F
× I
S
.
Figure 7.
An error budget illustrating the importance of low amplifier
input current, voltage offset, and offset voltage drift to minimize
output voltage errors can be developed by considering the equi-
valent circuit for the small (0.2 mm
2
area) photodiode shown in
Figure 7. The input current results in an error proportional to
the feedback resistance used. The amplifier’s offset will produce
an error proportional to the preamp’s noise gain (I + R
F
/R
SH
),
where R
SH
is the photodiode shunt resistance. The amplifier’s
input current will double with every 10°C rise in temperature,
and the photodiode’s shunt resistance halves with every 10°C
rise. The error budget in Figure 8 assumes a room temperature
photodiode R
SH
of 500 MΩ, and the maximum input current
and input offset voltage specs of an AD548C.
TEMP
CR
SH
(M)V
OS
(V) (1+ R
F
/R
SH
) V
OS
I
B
(pA) I
B
R
F
TOTAL
–
25 15,970 150 151 µV0.3030 µV 181 µV
0 2,830 200 207 µV2.26 262 µV 469 µV
25 500 250 300 µV 10.00 1.0 mV 1.30 mV
50 88.5 300 640 µV 56.6 5.6 mV 6.24 mV
75 15.6 350 2.6 mV 320 32 mV 34.6 mV
85 7.8 370 5.1 mV 640 64 mV 69.1 mV
Figure 8. Photodiode Preamp Errors Over Temperature
The capacitance at the amplifier’s negative input (the sum of the
photodiode’s shunt capacitance, the op amp’s differential input
capacitance, stray capacitance due to wiring, etc.) will cause a
rise in the preamp’s noise gain over frequency. This can result in
excess noise over the bandwidth of interest. C
F
reduces the
noise gain “peaking” at the expense of bandwidth.
INSTRUMENTATION AMPLIFIER
The AD548C’s maximum input current of 10 pA makes it an
excellent building block for the high input impedance instru-
mentation amplifier shown in Figure 9. Total current drain for
this circuit is under 600 µA. This configuration is optimal for
conditioning differential voltages from high impedance sources.
The overall gain of the circuit is controlled by R
G
, resulting in
the following transfer function:
V
OUT
V
IN
= 1 +
(R
1
+ R
2
)
R
G
REV. D
–9–
