MAX1953/MAX1954/MAX1957
Low-Cost, High-Frequency, Current-Mode PWM
Buck Controller
18 ______________________________________________________________________________________
The MAX1953/MAX1954/MAX1957s’ response to a load
transient depends on the selected output capacitors. In
general, more low-ESR output capacitance results in
better transient response. After a load transient, the
output voltage instantly changes by ESR ✕ ∆I
LOAD
.
Before the controller can respond, the output voltage
deviates further, depending on the inductor and output
capacitor values. After a short period of time (see the
Typical Operating Characteristics), the controller
responds by regulating the output voltage back to its
nominal state. The controller response time depends on
its closed-loop bandwidth. With a higher bandwidth,
the response time is faster, preventing the output volt-
age from further deviation from its regulating value.
Compensation Design
The MAX1953/MAX1954/MAX1957 use an internal
transconductance error amplifier whose output com-
pensates the control loop. The external inductor, high-
side MOSFET, output capacitor, compensation resistor,
and compensation capacitors determine the loop sta-
bility. The inductor and output capacitors are chosen
based on performance, size, and cost. Additionally, the
compensation resistor and capacitors are selected to
optimize control-loop stability. The component values
shown in the Typical Application Circuits (Figures 1
through 4) yield stable operation over the given range
of input-to-output voltages and load currents.
The controller uses a current-mode control scheme that
regulates the output voltage by forcing the required
current through the external inductor. The MAX1953/
MAX1954/MAX1957 use the voltage across the high-
side MOSFET’s on-resistance (R
DS(ON)
) to sense the
inductor current. Current-mode control eliminates the
double pole in the feedback loop caused by the induc-
tor and output capacitor, resulting in a smaller phase
shift and requiring less elaborate error-amplifier com-
pensation. A simple single-series R
C
and C
C
is all that
is needed to have a stable high bandwidth loop in
applications where ceramic capacitors are used for
output filtering. For other types of capacitors, due to the
higher capacitance and ESR, the frequency of the zero
created by the capacitance and ESR is lower than the
desired close loop crossover frequency. Another com-
pensation capacitor should be added to cancel this
ESR zero.
The basic regulator loop may be thought of as a power
modulator, output feedback divider, and an error ampli-
fier. The power modulator has DC gain set by g
mc
x
R
LOAD
, with a pole and zero pair set by R
LOAD
, the out-
put capacitor (C
OUT
), and its equivalent series resis-
tance (R
ESR
).
Below are equations that define the power modulator:
where R
LOAD
= V
OUT
/I
OUT(MAX)
, and g
mc
= 1/(A
CS
✕
R
DS(ON)
), where A
CS
is the gain of the current-sense
amplifier and R
DS(ON)
is the on-resistance of the high-
side power MOSFET. A
CS
is 6.3 for the MAX1953 when
ILIM is connected to GND, and 3.5 for the MAX1954/
MAX1957 and for the MAX1953 when ILIM is connect-
ed to V
IN
or floating. The frequencies at which the pole
and zero due to the power modulator occur are deter-
mined as follows:
The feedback voltage-divider used has a gain of G
FB
=
V
FB
/V
OUT
, where V
FB
is equal to 0.8V. The transcon-
ductance error amplifier has DC gain, G
EA(DC)
= gm ✕
R
O
. R
O
is typically 10MΩ. A dominant pole is set by the
compensation capacitor (C
C
), the amplifier output
resistance (R
O
), and the compensation resistor (R
C
). A
zero is set by the compensation resistor (R
C
) and the
compensation capacitor (C
C
).
There is an optional pole set by C
f
and R
C
to cancel the
output capacitor ESR zero if it occurs before crossover
frequency (f
C
):
The crossover frequency (f
C
) should be much higher
than the power modulator pole f
pMOD
. Also, the
crossover frequency should be less than 1/5 the
switching frequency:
