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LTC3729EUH#TRPBF

P1-P3P4-P6P7-P9P10-P12P13-P15P16-P18P19-P21P22-P24P25-P27P28-P30
LTC3729
13
3729fb
Accepting larger values of ∆I
L
allows the use of low in‑
ductances, but can result in higher output voltage ripple.
A reasonable starting point for setting ripple current is
∆I
L
= 0.4(I
OUT
)/N, where N is the number of channels and
I
OUT
is the total load current. Remember, the maximum
∆I
L
occurs at the maximum input voltage. The individual
inductor ripple currents are constant determined by the
inductor, input and output voltages.
Inductor Core Selection
Once the values for L1 and L2 are known, the type of
inductor must be selected. High efficiency converters
generally cannot afford the core loss found in low cost
powdered iron cores, forcing the use of more expensive
ferrite, molypermalloy, or Kool Mµ
®
cores. Actual core loss
is independent of core size for a fixed inductor value, but it
is very dependent on inductance selected. As inductance
increases, core losses go down. Unfortunately, increased
inductance requires more turns of wire and therefore cop‑
per losses will increase.
Ferrite designs have very low core loss and are preferred
at high switching frequencies, so design goals can con‑
centrate on copper loss and preventing saturation. Ferrite
core material saturates “hard,” which means that induc‑
tance collapses abruptly when the peak design current is
exceeded. This results in an abrupt increase in inductor
APPLICATIONS INFORMATION
ripple current and consequent output voltage ripple. Do
not allow the core to saturate!
Molypermalloy (from Magnetics, Inc.) is a very good, low
loss core material for toroids, but it is more expensive
than ferrite. A reasonable compromise from the same
manufacturer is Kool Mµ. Toroids are very space effi‑
cient, especially when you can use several layers of wire.
Because they lack a bobbin, mounting is more difficult.
However, designs for surface mount are available which
do not increase the height significantly.
Power MOSFET, D1 and D2 Selection
Two external power MOSFETs must be selected for each
controller with the LTC3729: One N‑channel MOSFET for
the top (main) switch, and one N‑channel MOSFET for the
bottom (synchronous) switch.
The peak‑to‑peak drive levels are set by the INTV
CC
voltage.
This voltage is typically 5V during start‑up (see EXTV
CC
Pin
Connection). Consequently, logic‑level threshold MOSFETs
must be used in most applications. The only exception
is if low input voltage is expected (V
IN
< 5V); then, sub‑
logic‑level threshold MOSFETs (V
GS(TH)
< 3V) should be
used. Pay close attention to the BV
DSS
specification for
the MOSFETs as well; most of the logic‑level MOSFETs
are limited to 30V or less.
Selection criteria for the power MOSFETs include the
“ON” resistance R
DS(ON)
, reverse transfer capacitance
C
RSS
, input voltage, and maximum output current. When
the LTC3729 is operating in continuous mode the duty
factors for the top and bottom MOSFETs of each output
stage are given by:
Main Switch Duty Cycle =
V
OUT
V
IN
Synchronous Switch Duty Cycle =
V
IN
– V
OUT
V
IN
The MOSFET power dissipations at maximum output
current are given by:
Kool Mµ is a registered trademark of Magnetics, Inc.
Figure 3. Normalized Peak Output Current vs
Duty Factor [I
RMS
≈ 0.3 (∆I
O(P–P)
)]
DUTY FACTOR (V
OUT
/V
IN
)
0.1 0.2 0.3 0.4
0.5 0.6 0.7 0.8 0.9
1.0
0.9
0.8
0.7
0.6
0.5
0.4
0.3
0.2
0.1
0
3729 F03
6-PHASE
4-PHASE
3-PHASE
2-PHASE
1-PHASE
∆I
O(P-P)
V
O
/fL
LTC3729
14
3729fb
P
MAIN
=
V
OUT
V
IN
I
MAX
N
2
1+ d
( )
R
DS(ON)
+
k V
IN
( )
2
I
MAX
N
C
RSS
( )
f
( )
P
SYNC
=
V
IN
– V
OUT
V
IN
I
MAX
N
2
1+ d
( )
R
DS(ON)
where d is the temperature dependency of R
DS(ON)
, k is a
constant inversely related to the gate drive current and N
is the number of stages.
Both MOSFETs have I
2
R losses but the topside N‑channel
equation includes an additional term for transition losses,
which peak at the highest input voltage. For V
IN
< 20V
the high current efficiency generally improves with larger
MOSFETs, while for V
IN
> 20V the transition losses rapidly
increase to the point that the use of a higher R
DS(ON)
device
with lower C
RSS
actual provides higher efficiency. The
synchronous MOSFET losses are greatest at high input
voltage when the top switch duty factor is low or during
a short‑circuit when the synchronous switch is on close
to 100% of the period.
The term (1 + d) is generally given for a MOSFET in the
form of a normalized R
DS(ON)
vs. Temperature curve,
but d = 0.005/°C can be used as an approximation for
low voltage MOSFETs. C
RSS
is usually specified in the
MOSFET characteristics. The constant k = 1.7 can be used
to estimate the contributions of the two terms in the main
switch dissipation equation.
The Schottky diodes, D1 and D2 shown in Figure 1 conduct
during the dead‑time between the conduction of the two large
power MOSFETs. This helps prevent the body diode of the
bottom MOSFET from turning on, storing charge during the
dead‑time, and requiring a reverse recovery period which
would reduce efficiency. A 1A to 3A (depending on output
current) Schottky diode is generally a good compromise for
both regions of operation due to the relatively small average
current. Larger diodes result in additional transition losses
due to their larger junction capacitance.
APPLICATIONS INFORMATION
C
IN
and C
OUT
Selection
In continuous mode, the source current of each top
N‑channel MOSFET is a square wave of duty cycle
V
OUT
/V
IN
. A low ESR input capacitor sized for the maximum
RMS current must be used. The details of a close form
equation can be found in Application Note 77. Figure 4
shows the input capacitor ripple current for different phase
configurations with the output voltage fixed and input volt‑
age varied. The input ripple current is normalized against
the DC output current. The graph can be used in place of
tedious calculations. The minimum input ripple current
can be achieved when the product of phase number and
output voltage, N(V
OUT
), is approximately equal to the
input voltage V
IN
or:
V
OUT
V
IN
=
k
N
where k = 1, 2, …, N – 1
So the phase number can be chosen to minimize the input
capacitor size for the given input and output voltages.
In the graph of Figure 4, the local maximum input RMS
capacitor currents are reached when:
V
OUT
V
IN
=
2k − 1
2N
where k = 1, 2, …, N
These worst‑case conditions are commonly used for
design because even significant deviations do not offer
much relief. Note that capacitor manufacturer’s ripple
current ratings are often based on only 2000 hours of life.
Figure 4. Normalized Input RMS Ripple Current vs
Duty Factor for 1 to 6 Output Stages
DUTY FACTOR (V
OUT
/V
IN
)
0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8
0.9
0.6
0.5
0.4
0.3
0.2
0.1
0
3729 F04
RMS INPUT RIPPLE CURRNET
DC LOAD CURRENT
6-PHASE
4-PHASE
3-PHASE
2-PHASE
1-PHASE
LTC3729
15
3729fb
This makes it advisable to further derate the capacitor, or
to choose a capacitor rated at a higher temperature than
required. Several capacitors may also be paralleled to meet
size or height requirements in the design. Always consult
the capacitor manufacturer if there is any question.
The graph shows that the peak RMS input current is
reduced linearly, inversely proportional to the number, N
of stages used. It is important to note that the efficiency
loss is proportional to the input RMS current squared
and therefore a 2‑stage implementation results in 75%
less power loss when compared to a single phase design.
Battery/input protection fuse resistance (if used), PC board
trace and connector resistance losses are also reduced by
the reduction of the input ripple current in a PolyPhase
system. The required amount of input capacitance is further
reduced by the factor, N, due to the effective increase in
the frequency of the current pulses.
The selection of C
OUT
is driven by the required effective
series resistance (ESR). Typically once the ESR require‑
ment has been met, the RMS current rating generally far
exceeds the I
RIPPLE(P‑P)
requirements. The steady state
output ripple (∆V
OUT
) is determined by:
∆V
OUT
≈ ∆I
RIPPLE
ESR+
1
8NfC
OUT
Where f = operating frequency of each stage, N is the
number of phases, C
OUT
= output capacitance, and
∆I
RIPPLE
= combined inductor ripple currents.
The output ripple varies with input voltage since ∆I
L
is a
function of input voltage. The output ripple will be less than
50mV at max V
IN
with ∆I
L
= 0.4I
OUT(MAX)
/N assuming:
C
OUT
required ESR < 2N(R
SENSE
) and
C
OUT
> 1/(8Nf)(R
SENSE
)
The emergence of very low ESR capacitors in small,
surface mount packages makes very physically small
implementations possible. The ability to externally
compensate the switching regulator loop using the
I
TH
pin(OPTI‑LOOP compensation) allows a much
wider selection of output capacitor types. OPTI‑LOOP
compensation effectively removes constraints on output
capacitor ESR. The impedance characteristics of each
APPLICATIONS INFORMATION
capacitor type are significantly different than an ideal
capacitor and therefore require accurate modeling or
bench evaluation during design.
Manufacturers such as Nichicon, United Chemicon and
Sanyo should be considered for high performance through‑
hole capacitors. The OS‑CON semiconductor dielectric
capacitor available from Sanyo and the Panasonic SP
surface mount types have the lowest (ESR)(size) product
of any aluminum electrolytic at a somewhat higher price.
An additional ceramic capacitor in parallel with OS‑CON
type capacitors is recommended to reduce the inductance
effects.
In surface mount applications, multiple capacitors may
have to be paralleled to meet the ESR or RMS current
handling requirements of the application. Aluminum
electrolytic and dry tantalum capacitors are both available
in surface mount configurations. New special polymer
surface mount capacitors offer very low ESR also but
have much lower capacitive density per unit volume. In
the case of tantalum, it is critical that the capacitors are
surge tested for use in switching power supplies. Several
excellent choices are the AVX TPS, AVX TPSV or the KEMET
T510 series of surface mount tantalums, available in case
heights ranging from 2mm to 4mm. Other capacitor types
include Sanyo OS‑CON, Nichicon PL series and Sprague
595D series. Consult the manufacturer for other specific
recommendations. A combination of capacitors will often
result in maximizing performance and minimizing overall
cost and size.
INTV
CC
Regulator
An internal P‑channel low dropout regulator produces
5V at the INTV
CC
pin from the V
IN
supply pin. The INTV
CC
regulator powers the drivers and internal circuitry of the
LTC3729. The INTV
CC
pin regulator can supply up to
50mA peak and must be bypassed to power ground with
a minimum of 4.7µF tantalum or electrolytic capacitor. An
additional 1µF ceramic capacitor placed very close to the IC
is recommended due to the extremely high instantaneous
currents required by the MOSFET gate drivers.
High input voltage applications in which large MOSFETs
are being driven at high frequencies may cause the
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LTC3729EUH#TRPBF

Mfr. #:
Buy LTC3729EUH#TRPBF
Manufacturer:
Analog Devices / Linear Technology
Description:
Switching Voltage Regulators PolyPhase Controller QFN Package
Lifecycle:
New from this manufacturer.
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