MAX8650
4.5V to 28V Input Current-Mode Step-Down
Controller with Adjustable Frequency
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where I
P-P
is the peak-to-peak inductor current:
These equations are suitable for initial capacitor selec-
tion, but final values should be chosen based on a proto-
type or evaluation circuit. As a general rule, a smaller
current ripple results in less output-voltage ripple. Since
the inductor ripple current is a factor of the inductor
value and input voltage, the output-voltage ripple
decreases with larger inductance, and increases with
higher input voltages. Ceramic, tantalum, or aluminum
polymer electrolytic capacitors are recommended. The
aluminum electrolytic capacitor is the least expensive;
however, it has higher ESR. To compensate for this, use
a ceramic capacitor in parallel to reduce the switching
ripple and noise. For reliable and safe operation, ensure
that the capacitor’s voltage and ripple-current ratings
exceed the calculated values.
The response to a load transient depends on the
selected output capacitors. After a load transient, the
output voltage instantly changes by ESR x ΔI
LOAD
.
Before the controller can respond, the output voltage
deviates further, depending on the inductor and output-
capacitor values. After a short period (see the
Typical
Operating Characteristics
), the controller responds by
regulating the output voltage back to its nominal state.
The controller response time depends on its closed-
loop bandwidth. With a higher bandwidth, the response
time is faster, thus preventing the output voltage from
further deviation from its regulating value.
Compensation Design
The MAX8650 uses an internal transconductance error
amplifier whose output compensates the control loop.
The external inductor, output capacitor, compensation
resistor, and compensation capacitors determine the
loop stability. The inductor and output capacitor are
chosen based on performance, size, and cost.
Additionally, the compensation resistor and capacitors
are selected to optimize control-loop stability. The com-
ponent values, shown in the circuits of Figures 3 and 4,
yield stable operation over the given range of input-to-
output voltages.
The controller uses a current-mode control scheme that
regulates the output voltage by forcing the required cur-
rent through the external inductor, so the MAX8650 uses
the voltage drop across the DC resistance of the induc-
tor or the alternate series current-sense resistor to mea-
sure the inductor current. Current-mode control elimi-
nates the double pole in the feedback loop caused by
the inductor and output capacitor resulting in a smaller
phase shift and requiring a less elaborate error-amplifier
compensation than voltage-mode control. A simple sin-
gle-series R
C
and C
C
is all that is needed to have a sta-
ble, high-bandwidth loop in applications where ceramic
capacitors are used for output filtering. For other types
of capacitors, due to the higher capacitance and ESR,
the frequency of the zero created by the capacitance
and ESR is lower than the desired closed-loop
crossover frequency. To stabilize a nonceramic output
capacitor loop, add another compensation capacitor
from COMP to GND to cancel this ESR zero.
The basic regulator loop is modeled as a power modu-
lator, an output feedback-divider, and an error amplifi-
er. The power modulator has DC gain set by g
mc
x
R
LOAD
, with a pole and zero pair set by R
LOAD
, the out-
put capacitor (C
OUT
), and its ESR. Below are equations
that define the power modulator:
where R
LOAD
= V
OUT
/ I
OUT(MAX)
, f
S
is the switching
frequency, L is the output inductance, and g
mc
= 1 /
(A
VCS
x R
DC
), where A
VCS
is the gain of the current-
sense amplifier (12 typ), and R
DC
is the DC resistance
of the inductor.
Find the pole and zero frequencies created by the
power modulator as follows:
When C
OUT
comprises “n” identical capacitors in paral-
lel, the resulting C
OUT
= n x C
OUT(EACH)
, and ESR =
ESR
(EACH)
/ n. Note that the capacitor zero for a paral-
lel combination of like capacitors is the same as for an
individual capacitor. See Figures 10 and 11 for illustra-
tions of the pole and zero locations.
The feedback voltage-divider has a gain of G
FB
= V
FB
/
V
OUT
, where V
FB
is equal to 0.75V.
