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LT3988HMSE#TRPBF

P1-P3P4-P6P7-P9P10-P12P13-P15P16-P18P19-P21P22-P24
LT3988
13
3988f
applicaTions inForMaTion
The boost circuit also limits the minimum input voltage for
proper start-up. If the input voltage ramps slowly, or the
LT3988 turns on when the output is already in regulation,
the boost capacitor may not be fully charged. Because
the boost capacitor charges with the energy stored in the
inductor, the circuit will rely on some minimum load current
to get the boost circuit running properly. This minimum
load will depend on input and output voltages, and on the
arrangement of the boost circuit. The minimum load cur-
rent generally goes to zero once the circuit has started.
Figure 4 shows a plot of input voltage to start and to run
as a function of load current. Even without an output load
current, in many cases the discharged output capacitor
will present a load to the switcher that will allow it to start.
The boost current is generally small but can become sig-
nificant at high duty cycles. The required boost current is:
I
BOOST
=
V
OUT
V
IN
I
OUT
40
Figure 3. Diode D4 Prevents a Shorted Input from
Discharging a Backup Battery Tied to the Output
Figure 4. The Minimum Input Voltage Depends on Output Voltage, Load Current, and Boost Circuit
Figure 2. Generating the Boost Voltage
Converter with Backup Output Regulator
There is another situation to consider: systems where the
output will be held high when the input to the LT3988 is
absent. If the V
IN
pin is grounded while the output is held
high, then diodes inside the LT3988 can pull large currents
from the output through the SW and V
IN
pins. A Schottky
diode in series with the input to the LT3988, as shown in
Figure 3, will protect the LT3988 and the system from a
shorted or reversed input.
V
IN
V
OUT
V
BOOST
– V
SW
≅ V
OUT
MAX V
BOOST
≅ V
IN
+ V
OUT
3988 F02
C3
V
IN
SW
GND
BD BOOST
(2a)
V
IN
V
OUT
V
BOOST
– V
SW
≅ V
IN
MAX V
BOOST
≅ 2V
IN
C3
V
IN
SW
GND
BD BOOST
(2b)
V
IN
V
OUT
V
BOOST
– V
SW
≅ V
IN3
MAX V
BOOST
≅ V
IN3
+ V
IN
MIN VALUE FOR V
IN3
= 3V
V
IN
SW
GND
BD BOOST
(2c)
V
IN3
> 3V
3988 F03
V
IN
D4
SW
GND
V
OUT
LT3988
LOAD CURRENT (mA)
0
5.0
5.5
800
3988 F04a
4.5
4.0
400200 600 1000
3.5
INPUT VOLTAGE (V)
T
A
= 25°C
TO START
TO RUN
LOAD CURRENT (mA)
0
6.6
7.0
800
3988 F04b
6.2
5.8
400200 600 1000
5.4
INPUT VOLTAGE (V)
T
A
= 25°C
TO START
TO RUN
Minimum Input Voltage, V
OUT
= 3.3V Minimum Input Voltage, V
OUT
= 5V
LT3988
14
3988f
applicaTions inForMaTion
Input Capacitor Selection
Bypass the input of the LT3988 circuit with a 4.7μF or higher
ceramic capacitor of X7R or X5R type. A lower value or
a less expensive Y5V type will work if there is additional
bypassing provided by bulk electrolytic capacitors, or if the
input source impedance is low. The following paragraphs
describe the input capacitor considerations in more detail.
Step-down regulators draw current from the input supply
in pulses with very fast rise and fall times. The input ca-
pacitor is required to reduce the resulting voltage ripple at
the LT3988 input and to force this switching current into a
tight local loop, minimizing EMI. The input capacitor must
have low impedance at the switching frequency to do this
effectively and it must have an adequate ripple current rat-
ing. With two switchers operating at the same frequency
but with different phases and duty cycles, calculating the
input capacitor RMS current is not simple; however, a
conservative value is the RMS input current for the phase
delivering the most power (V
OUT
• I
OUT
):
I
IN(RMS)
= I
OUT
•
V
OUT
V
IN
– V
OUT
( )
V
IN
<
I
OUT
2
and is largest when V
IN
= 2V
OUT
(50% duty cycle). As
the second, lower power channel draws input current,
the input capacitor’s RMS current actually decreases as
the out-of-phase current cancels the current drawn by the
higher power channel. Considering that the maximum load
current from a single phase (if SW1 and SW2 are both at
maximum current) is ~1A, RMS ripple current will always
be less than 0.5A.
The high frequency of the LT3988 reduces the energy
storage requirements of the input capacitor, so that the
capacitance required is often less than 10μF. The combi-
nation of small size and low impedance (low equivalent
series resistance or ESR) of ceramic capacitors makes
them the preferred choice. The low ESR results in very
low voltage ripple. Ceramic capacitors can handle larger
magnitudes of ripple current than other capacitor types
of the same value.
An alternative to a high value ceramic capacitor is a lower
value along with a larger electrolytic capacitor, for example
a 1μF ceramic capacitor in parallel with a low ESR tantalum
capacitor. For the electrolytic capacitor, a value larger than
10μF will be required to meet the ESR and ripple current
requirements. Because the input capacitor is likely to see
high surge currents when the input source is applied, tan-
talum capacitors should be surge rated. The manufacturer
may also recommend operation below the rated voltage
of the capacitor. Be sure to place the 1μF ceramic as close
as possible to the V
IN
and GND pins on the IC for optimal
noise immunity.
A final caution is in order regarding the use of ceramic
capacitors at the input. A ceramic input capacitor can
combine with stray inductance to form a resonant tank
circuit. If power is applied quickly (for example by plugging
the circuit into a live power source), this tank can ring,
doubling the input voltage and damaging the LT3988. The
solution is to either clamp the input voltage or dampen the
tank circuit by adding a lossy capacitor in parallel with the
ceramic capacitor. For details, see Application Note 88.
Frequency Compensation
The LT3988 uses current mode control to regulate the
output. This simplifies loop compensation. In particular, the
LT3988 does not depend on the ESR of the output capacitor
for stability, so you are free to use ceramic capacitors to
achieve low output ripple and small circuit size. The LT3988
is internally compensated with the RC network tied to the
VC node. The internal compensation network is optimized
to provide stability over the full frequency range. Figure 5
shows an equivalent circuit for the LT3988 control loop.
The error amplifier is a transconductance amplifier with
0.75V
LT3988
3988 F05
R1
OUT
R
ESR
C
C
40pF
R
C
300k
V
C
7M
ERROR
AMPLIFIER
FB
R2
C
OUT
CURRENT MODE
POWER STAGE
C
PL
g
m
= 2A/V
g
m
= 40µA/V
Figure 5. Model For Loop Response
LT3988
15
3988f
Output Voltage Tracking
The LT3988 allows the user to program how the output
ramps up by means of the TRACK/SS pins. Through these
pins, either channel output can be set up to either coin-
cidently or ratiometrically track the other channel output.
This example will show the channel 2 output tracking the
channel 1 output, as shown in Figure 7.
The TRACK/SS2 pin acts as a clamp on channel 2’s ref-
erence voltage. V
OUT2
is referenced to the TRACK/SS2
voltage when the TRACK/SS2 < 0.8V and to the internal
precision reference when TRACK/SS2 > 0.8V. To imple-
ment the coincident tracking in Figure 7, connect an extra
resistive divider to the output of channel 1 and connect its
midpoint to the TRACK/SS2 pin (Figure 8).
The ratio of this divider should be selected to be the
same as that of channel 2’s feedback divider (R5 = R3
and R6 = R4). In this tracking mode, V
OUT1
must be set
higher than V
OUT2
. To implement the ratiometric tracking
in Figure 6, change the extra divider ratio to R5 = R1 and
R6 = R2 + ∆R. The extra resistance on R6 should be set
so that the TRACK/SS2 voltage is ≥1V when V
OUT1
is at
its final value. The need for this extra resistance is best
understood with the help of the equivalent input circuit
shown in Figure 9.
applicaTions inForMaTion
finite output impedance. The power section, consisting of
the modulator, power switch and inductor, is modeled as a
transconductance amplifier generating an output current
proportional to the voltage at the V
C
node.
Note that the output capacitor integrates this current, and
that the capacitor on the V
C
node (C
C
) integrates the er-
ror amplifier output current, resulting in two poles in the
loop. R
C
provides a zero. With the recommended output
capacitor, the loop crossover occurs above the R
C
C
C
zero.
This simple model works well as long as the value of the
inductor is not too high and the loop crossover frequency
is much lower than the switching frequency. With a larger
ceramic capacitor (very low ESR), crossover may be lower
and a phase lead capacitor (CPL) across the feedback
divider may improve the phase margin and transient
response. Large electrolytic capacitors may have an ESR
large enough to create an additional zero, and the phase
lead may not be necessary. If the output capacitor is dif-
ferent than the recommended capacitor, stability should
be checked across all operating conditions, including
load current, input voltage and temperature. The LT1375
data sheet contains a more thorough discussion of loop
compensation and describes how to test the stability us-
ing a transient load.
Shutdown
The EN/UVLO pin is used for two purposes, to place the
LT3988 in a low current shutdown mode, and to override
the internal undervoltage lockout thresholds with a user
programmable threshold. When the EN/UVLO pin is pulled
to under 0.5V (typ), the LT3988 is in shutdown mode and
draws less than 1µA from the input supply. When the
EN/UVLO pin is driven above 0.5V (typ) and less than 1.2V
(typ), the internal regulator is activated and the oscillators
are operating, but the switching operation of both chan-
nels remains inhibited. When the EV/UVLO pin is driven
above 1.2V (typ), the undervoltage lockout asserted by the
EN/UVLO function is released, allowing switching opera-
tion of both channels. Internal undervoltage detectors will
still prevent switching operation on channel 1 until V
IN1
is
greater than 3.9V (typ) and on channel 2 until V
IN2
is greater
than 2.6V (typ). The EN/UVLO undervoltage lockout has
120mV (typ) of hysteresis. The EN/UVLO pin is rated up
to 80V and can be connected directly to the input voltage.
The EN/UVLO pin may be driven by a voltage divider from
V
IN1
, allowing an externally programmable undervoltage
lockout to be set above the internal 3.9V threshold. The
undervoltage threshold and hysteresis are given by:
V
UVTH
= 1.2 1+
R1
R2
;R1= R2
V
UVTH
1.2
–1
V
UVHY
= 0.12 1+
R1
R2
;R1= R2
V
UVHY
0.12
–1
R2
V
IN1
3988 F06
R1
EN/UVLO
1.2V
UVLO
+
–
Figure 6. Undervoltage Lockout Circuit
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LT3988HMSE#TRPBF

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Buy LT3988HMSE#TRPBF
Manufacturer:
Analog Devices / Linear Technology
Description:
Switching Voltage Regulators Dual 60V Monolithic 1A Step-Down Switching Regulator
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