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LTC3408EDD#TRPBF

P1-P3P4-P6P7-P9P10-P12
7
LTC34 08
3408f
OPERATIO
U
(Refer to Functional Diagram)
off and on the bypass P-channel MOSFET with a frequency
of approximately 50kHz to 100kHz at 1.6A peak current.
This will continue until the short is removed. While the
bypass P-channel MOSFET is pulsing intermittently, the
inherent current limit of the step-down regulator limits its
peak current to about 1A.
Dropout Operation
If the reference voltage would cause V
OUT
to exceed V
IN
,
the LTC3408 enters dropout operation. During dropout,
the main switch remains on continuously and operates at
100% duty cycle. If the voltage at REF is less than 1.2V, the
bypass P-channel MOSFET will stay off even in dropout
operation. The output voltage is then determined by the
input voltage minus the voltage drop across the main switch
and the inductor. If the voltage at REF is greater than 1.2V,
Figure 2. Maximum Output Current vs Input Voltage
SUPPLY VOLTAGE (V)
2.5
MAXIMUM OUTPUT CURRENT (mA)
1200
1000
800
600
400
200
0
3.0
3.5 4.0 4.5
3408 F02
5.0 5.5
V
OUT
= 1.8V
V
OUT
= 1.5V
V
OUT
= 2.5V
but less than V
IN
/3, the bypass P-channel MOSFET will be
on, but the main switch will be off. For best performance
and lowest voltage drop from V
IN
to V
OUT
, always ensure
that the REF voltage is greater than both 1.2V and V
IN
/3.
An important detail to remember is that at low input
supply voltages, the R
DS(ON)
of the P-channel switch
increases (see Typical Performance Characteristics).
Therefore, the user should calculate the power dissipa-
tion when the LTC3408 is used at 100% duty cycle with
low input voltage (See Thermal Considerations in the
Applications Information section).
Low Supply Operation
The LTC3408 will operate with input supply voltages as
low as 2.5V, but the maximum allowable output current is
reduced at this low voltage. Figure 2 shows the reduction
in the maximum output current as a function of input
voltage for various output voltages.
Slope Compensation and Inductor Peak Current
Slope compensation provides stability in constant fre-
quency architectures by preventing subharmonic oscilla-
tions at high duty cycles. It is accomplished internally by
adding a compensating ramp to the inductor current
signal at duty cycles in excess of 40%. Normally, this
results in a reduction of maximum inductor peak current
for duty cycles >40%. However, the LTC3408 uses a
patent-pending scheme that counteracts this compensat-
ing ramp, which allows the maximum inductor peak
current to remain unaffected throughout all duty cycles.
APPLICATIO S I FOR ATIO
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The basic LTC3408 application circuit is shown in Fig-
ure 1. External component selection is driven by the load
requirement and begins with the selection of L followed by
C
IN
and C
OUT
.
Inductor Selection
For most applications, the value of the inductor will fall in
the range of 4µH to 6µH. Its value is chosen based on the
desired ripple current. Large value inductors lower ripple
current and small value inductors result in higher ripple
currents. As Equation 1 shows, a greater difference be-
tween V
IN
and V
OUT
produces a larger ripple current.
Where these voltages are subject to change, the highest
V
IN
and lowest V
OUT
will determine the maximum ripple
current. A reasonable starting point for setting ripple
current is I
L
= 120mA (20% of the maximum load, 600mA).
∆=
I
fL
V
V
V
L OUT
OUT
IN
1
1
()()
–
(1)
8
LTC34 08
3408f
At output voltages below 0.6V, the switching frequency
decreases linearly to a minimum of approximately 700kHz.
This places the maximum ripple current (in forced con-
tinuous mode) at the highest input voltage and the lowest
output voltage. In practice, the resulting ouput ripple
voltage is 10mV to 15mV using the components specified
in Figure 1.
The DC current rating of the inductor should be at least equal
to the maximum load current plus half the ripple current to
prevent core saturation. Thus, a 660mA rated inductor
should be enough for most applications (600mA + 60mA).
For better efficiency, choose a low DC-resistance inductor.
Inductor Core Selection
Different core materials and shapes will change the size/
current and price/current relationship of an inductor.
Toroid or shielded pot cores in ferrite or permalloy mate-
rials are small and don’t radiate much energy but generally
cost more than powdered iron core inductors with similar
electrical characteristics. The choice of which style induc-
tor to use often depends more on the price versus size
requirements and any radiated field/EMI requirements
than on what the LTC3408 requires to operate. Table 1
shows some typical surface mount inductors that work
well in LTC3408 applications.
Table 1. Representative Surface Mount Inductors
PART VALUE DCR MAX DC SIZE
NUMBER (µH) (ΩMAX) CURRENT (A) WxLxH (mm
3
)
Sumida 4.7 0.135 0.5 3.2 x 3.2 x 1.2
CDRH2D11
Sumida 4.7 0.078 0.63 3.2 x 3.2 x 2.0
CDRH2D18/LD
Sumida 4.7 0.216 0.75 3.5 x 4.1 x 0.8
CMD4D06
Murata 4.7 0.150 0.65 2.5 x 3.2 x 2.0
LQH32C
Taiyo Yuden 4.7 0.250 0.210 1.6 x 2.0 x 1.6
LBLQ2016
Toko 4.7 0.20 0.79 3.6 x 3.6 x 1.2
D312C
C
IN
and C
OUT
Selection
In continuous mode, the source current of the top MOSFET
is a square wave of duty cycle V
OUT
/V
IN
. To prevent large
voltage transients, a low ESR input capacitor sized for the
APPLICATIO S I FOR ATIO
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maximum RMS current must be used. The maximum
RMS capacitor current is given by:
C required I I
VVV
V
IN RMS OMAX
OUT IN OUT
IN
≅
[(– )]
/12
This formula has a maximum at V
IN
= 2V
OUT
, where I
RMS
= I
OUT
/2. This simple worst-case condition is commonly
used for design because even significant deviations do not
offer much relief. Note that the capacitor manufacturer’s
ripple current ratings are often based on 2000 hours of life.
This makes it advisable to further derate the capacitor, or
choose a capacitor rated at a higher temperature than re-
quired. Always consult the manufacturer if there is any
question.
The selection of C
OUT
is driven by the required effective
series resistance (ESR). Typically, once the ESR
requirement for C
OUT
has been met, the RMS current
rating generally far exceeds the I
RIPPLE(P-P)
requirement.
The output ripple V
OUT
is determined by:
∆≅∆ +
VIESR
fC
OUT L
OUT
1
8
where f = operating frequency, C
OUT
= output capacitance
and I
L
= ripple current in the inductor. For a fixed output
voltage, the output ripple is highest at maximum input
voltage since I
L
increases with input voltage.
Aluminum electrolytic and dry tantalum capacitors are
both available in surface mount configurations. In the case
of tantalum, it is critical that the capacitors are surge tested
for use in switching power supplies. An excellent choice is
the AVX TPS series of surface mount tantalum. These are
specially constructed and tested for low ESR so they give
the lowest ESR for a given volume. Other capacitor types
include Sanyo POSCAP, Kemet T510 and T495 series, and
Sprague 593D and 595D series. Consult the manufacturer
for other specific recommendations.
The bulk capacitance values in Figure 1(a) (C
IN
= 10µF,
C
OUT
= 4.7µF) are tailored to mobile phone applications, in
which the output voltage is expected to slew quickly
according to the needs of the power amplifier. Holding the
output capacitor to 4.7µF facilitates rapid charging and
discharging. When the output voltage descends quickly in
9
LTC34 08
3408f
forced continuous mode, the LTC3408 will actually pull
current from the output until the command from V
REF
is
satisfied. On alternate half cyles, this current actually exits
the V
IN
terminal, potentially causing a rise in V
IN
and
forcing current into the battery. To prevent deterioration
of the battery, use sufficient bulk capacitance with low
ESR; at least 10µF is recommended.
Using Ceramic Input and Output Capacitors
Higher values, lower cost ceramic capacitors are now
becoming available in smaller case sizes. Their high ripple
current, high voltage rating and low ESR make them ideal
for switching regulator applications. Because the
LTC3408’s control loop does not depend on the output
capacitor’s ESR for stable operation, ceramic capacitors
can be used freely to achieve very low output ripple and
small circuit size.
However, care must be taken when ceramic capacitors are
used at the input and the output. When a ceramic capacitor
is used at the input and the power is supplied by a wall
adapter through long wires, a load step at the output can
induce ringing at the input, V
IN
. At best, this ringing can
couple to the output and be mistaken as loop instability. At
worst, a sudden inrush of current through the long wires
can potentially cause a voltage spike at V
IN
large enough
to damage the part.
When choosing the input and output ceramic capacitors,
choose the X5R or X7R dielectric formulations. These
dielectrics have the best temperature and voltage charac-
teristics of all the ceramics for a given value and size.
Ceramic capacitors of Y5V material are not recommended
because normal operating voltages cause their bulk ca-
pacitance to become much less than the nominal value.
Programming the Output Voltage With a DAC
The output voltage can be dynamically programmed to any
voltage from 0.3V to 3.5V with an external DAC driving the
REF pin. When the output is commanded low, the output
voltage descends quickly in forced continuous mode
pulling current from the output and transferring it to the
input. If the input is not connected to a low impedance
source capable of absorbing the energy, the input voltage
could rise above the absolute maximum voltage of the part
and get damaged. The faster V
OUT
is commanded low, the
higher is the voltage spike at the input. For best results,
ramp the REF pin from high to low as slow as the
application will allow. Avoid abrupt changes in voltage of
>0.2V/µs. If ramp control is unavailable, an RC filter with
a time constant of 10µs can be inserted between the REF
pin and the DAC as shown in Figure 3.
APPLICATIO S I FOR ATIO
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Figure 3. Filtering the REF Pin
LTC3408
REF
3408 F03
GND
DAC
10k
1000pF
Efficiency Considerations
The efficiency of a switching regulator is equal to the
output power divided by the input power times 100%. It is
often useful to analyze individual losses to determine what
is limiting the efficiency and which change would produce
the most improvement. Efficiency can be expressed as:
Efficiency = 100% – (L1 + L2 + L3 + ...)
where L1, L2, etc. are the individual losses as a percentage
of input power.
Although all dissipative elements in the circuit produce
losses, two main sources usually account for most of the
losses in LTC3408 circuits: V
IN
quiescent current and I
2
R
losses. The V
IN
quiescent current loss dominates the effi-
ciency loss at low load currents whereas the I
2
R loss domi-
nates the efficiency loss at medium to high load currents.
In a typical efficiency plot, the efficiency curve at low load
currents can be misleading since the actual power lost is
of little consequence as illustrated in Figure 4.
1. The V
IN
quiescent current consists of two components:
the DC bias current as given in the electrical characteris-
tics and the internal main switch and synchronous switch
gate charge currents. The gate charge current results
from switching the gate capacitance of the internal power
MOSFET switches. Each time the gate is switched from
high to low to high again, a packet of charge, dQ, moves
from V
IN
to ground. The resulting dQ/dt is typically larger
than the DC bias current. In continuous mode,
P1-P3P4-P6P7-P9P10-P12

LTC3408EDD#TRPBF

Mfr. #:
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Manufacturer:
Analog Devices / Linear Technology
Description:
Switching Voltage Regulators 600mA, 1.5MHz Synch Step-down Cvrtr
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