Data Sheet AD8203
Rev. D | Page 15 of 20
GAIN TRIM
Figure 45 shows a method for incremental gain trimming by
using a trim potentiometer and external resistor R
EXT
.
The following approximation is useful for small gain ranges:
ΔG ≈ (10 MΩ/R
EXT
)%
Thus, the adjustment range is ±2% for R
EXT
= 5 MΩ; ±10% for
R
EXT
= 1 MΩ, and so on.
5V
OUT
R
EXT
GAIN TRIM
20kΩ MIN
V
CM
V
DIFF
2
V
DIFF
2
NC = NO CONNECT
GND
NC
–IN
+IN
A1
+V
S
A2
OUT
AD8203
05013-018
Figure 45. Incremental Gain Trim
Internal Signal Overload Considerations
When configuring gain for values other than 14, the maximum
input voltage with respect to the supply voltage and ground
must be considered, since either the preamplifier or the output
buffer reaches its full-scale output (approximately V
S
− 0.2 V)
with large differential input voltages. The input of the AD8203
is limited to (V
S
− 0.2)/7 for overall gains ≤ 7, since the pre-
amplifier, with its fixed gain of ×7, reaches its full-scale output
before the output buffer. For gains greater than 7, the swing at
the buffer output reaches its full scale first and limits the
AD8203 input to (V
S
− 0.2)/G, where G is the overall gain.
LOW-PASS FILTERING
In many transducer applications, it is necessary to filter the
signal to remove spurious high frequency components, includ-
ing noise, or to extract the mean value of a fluctuating signal
with a peak-to-average ratio (PAR) greater than unity. For
example, a full-wave rectified sinusoid has a PAR of 1.57, a
raised cosine has a PAR of 2, and a half-wave sinusoid has a
PAR of 3.14. Signals having large spikes can have PARs of
10 or more.
When implementing a filter, the PAR should be considered so
that the output of the AD8203 preamplifier (A1) does not clip
before A2, since this nonlinearity would be averaged and appear
as an error at the output. To avoid this error, both amplifiers
should be made to clip at the same time. This condition is
achieved when the PAR is no greater than the gain of the sec-
ond amplifier (2 for the default configuration). For example, if a
PAR of 5 is expected, the gain of A2 should be increased to 5.
Low-pass filters can be implemented in several ways by using
the features provided by the AD8203. In the simplest case, a
single-pole filter (20 dB/decade) is formed when the output of
A1 is connected to the input of A2 via the internal 100 kΩ
resistor by strapping Pin 3, Pin 4, and a capacitor added from
this node to ground, as shown in Figure 46. If a resistor is added
across the capacitor to lower the gain, the corner frequency
increases; it should be calculated using the parallel sum of the
resistor and 100 kΩ.
5V
V
CM
V
DIFF
2
V
DIFF
2
NC = NO CONNECT
C
GND
NC
–IN
+IN
A1
+V
S
A2
OUT
AD8203
05013-019
OUTPUT
f
C
=
1
2πC10
5
C IN FARADS
Figure 46. Single-Pole, Low-Pass Filter Using the Internal 100 kΩ Resistor
If the gain is raised using a resistor, as shown in Figure 44, the
corner frequency is lowered by the same factor as the gain is
raised. Thus, using a resistor of 200 kΩ (for which the gain
would be doubled), the corner frequency is now 0.796 Hz µF
(0.039 µF for a 20 Hz corner frequency).
5V
V
CM
V
DIFF
2
V
DIFF
2
NC = NO CONNECT
C
GND
NC
–IN
+IN
A1
+V
S
A2
OUT
AD8203
005013-020
OUT
C
255kΩ
f
C
(Hz) = 1/C(µF)
Figure 47. 2-Pole, Low-Pass Filter
A 2-pole filter (with a roll-off of 40 dB/decade) can be implemented
using the connections shown in Figure 47. This is a Sallen-Key
form based on a ×2 amplifier. It is useful to remember that a 2-pole
filter with a corner frequency f
2
and a 1-pole filter with a corner at f
1
have the same attenuation at the frequency (f
2
2
/f
1
). The attenuation
at that frequency is 40 log (f
2
/f
1
), which is illustrated in Figure 48.
Using the standard resistor value shown and equal capacitors (see
Figure 47), the corner frequency is conveniently scaled at 1 Hz µF
(0.05 µF for a 20 Hz corner). A maximally flat response occurs
when the resistor is lowered to 196 kΩ and the scaling is then
1.145 Hz µF. The output offset is raised by approximately 5 mV
(equivalent to 250 µV at the input pins).
