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LT1766IGN-5#TRPBF

P1-P3P4-P6P7-P9P10-P12P13-P15P16-P18P19-P21P22-P24P25-P27P28-P30
LT1766/LT1766-5
13
1766fc
APPLICATIONS INFORMATION
the high side for discontinuous mode, so it can be used
for all conditions.
II
I
I
VVV
VfL
PEAK OUT
LP P
OUT
OUT IN OUT
IN
=+ =+
()( )
()( )()()
()
–
-
2
2
EMI
Decide if the design can tolerate an open core geometry like
a rod or barrel, which have high magnetic fi eld radiation,
or whether it needs a closed core like a toroid to prevent
EMI problems. This is a tough decision because the rods
or barrels are temptingly cheap and small and there are
no helpful guidelines to calculate when the magnetic fi eld
radiation will be a problem.
Additional Considerations
After making an initial choice, consider additional factors
such as core losses and second sourcing, etc. Use the
experts in Linear Technology’s Applications department
if you feel uncertain about the fi nal choice. They have
experience with a wide range of inductor types and can tell
you about the latest developments in low profi le, surface
mounting, etc.
Maximum Output Load Current
Maximum load current for a buck converter is limited
by the maximum switch current rating (I
P
). The current
rating for the LT1766 is 1.5A. Unlike most current mode
converters, the LT1766 maximum switch current limit
does not fall off at high duty cycles. Most current mode
converters suffer a drop off of peak switch current for
duty cycles above 50%. This is due to the effects of slope
compensation required to prevent subharmonic oscilla-
tions in current mode converters. (For detailed analysis,
see Application Note 19.)
The LT1766 is able to maintain peak switch current limit
over the full duty cycle range by using patented circuitry*
to cancel the effects of slope compensation on peak switch
current without affecting the frequency compensation it
provides.
Maximum load current would be equal to maximum switch
current
for an infi nitely large inductor
, but with fi nite
inductor size, maximum load current is reduced by one-
half peak-to-peak inductor current (I
LP-P
). The following
formula assumes continuous mode operation, implying
that the term on the right is less than one-half of I
P
.
I
OUT(MAX)
=
Continuous Mode
I–
I
2
= I
P
LP-P
P
−
+
()
−
()
()()( )
VVVVV
LfV
OUT F IN OUT F
IN
–
2
For V
OUT
= 5V, V
IN
= 8V, V
F(D1)
= 0.63V, f = 200kHz and
L = 20μH:
I
A
OUT MAX
()
−
=−
+
()
−
()
()()
()
=− =
15
5 0 63 8 5 0 63
2 20 10 200 10 8
15 021 129
63
.
.–.
••
.. .
Note that there is less load current available at the higher
input voltage because inductor ripple current increases.
At V
IN
= 15V, duty cycle is 33% and for the same set of
conditions:
I
A
OUT MAX()
.
.–.
••
.. .
=−
+
()
−
()
()()
()
=− =
−
15
5 0 63 15 5 0 63
2 20 10 200 10 15
15 044 106
63
To calculate actual peak switch current with a given set
of conditions, use:
II
I
VVVVV
LfV
SW PEAK
OUT
P
OUT
OUT F IN OUT F
IN
()
=+
=+
+−
()
()()( )
I
2
L-P
() –
2
Reduced Inductor Value and Discontinuous Mode
If the smallest inductor value is of most importance to a
converter design, in order to reduce inductor size/cost,
discontinuous mode may yield the smallest inductor solu-
tion. The maximum output load current in discontinuous
mode, however, must be calculated and is defi ned later
in this section.
*Patent # 6, 498, 466
LT1766/LT1766-5
14
1766fc
Discontinuous mode is entered when the output load
current is less than one-half of the inductor ripple current
(I
LP-P
). In this mode, inductor current falls to zero before
the next switch turn on (see Figure 8). Buck converters
will be in discontinuous mode for output load current
given by:
I
OUT
Discontinuous Mode
The inductor value in a buck converter is usually chosen
large enough to keep inductor ripple current (I
LP-P
) low;
this is done to minimize output ripple voltage and maximize
output load current. In the case of large inductor values,
as seen in the equation above, discontinuous mode will
be associated with light loads.
When choosing small inductor values, however, discon-
tinuous mode will occur at much higher output load cur-
rents. The limit to the smallest inductor value that can be
chosen is set by the LT1766 peak switch current (I
P
) and
the maximum output load current required, given by:
I
OUT(MAX)
Discontinuous Mode
Example: For V
IN
= 15V, V
OUT
= 5V, V
F
= 0.63V, f = 200kHz
and L = 10μH.
I
OUT(MAX)
Discontinuous
Mode
I
OUT(MAX)
= 0.639A
Discontinuous Mode
What has been shown here is that if high inductor ripple
current and discontinuous mode operation can be tolerated,
small inductor values can be used. If a higher output load
current is required, the inductor value must be increased.
If I
OUT(MAX)
no longer meets the discontinuous mode
criteria, use the I
OUT(MAX)
equation for continuous mode;
the LT1766 is designed to operate well in both modes of
operation, allowing a large range of inductor values to
be used.
Short-Circuit Considerations
The LT1766 is a current mode controller. It uses the V
C
node voltage as an input to a current comparator which
turns off the output switch on a cycle-by-cycle basis as
this peak current is reached. The internal clamp on the V
C
node, nominally 2V, then acts as an output switch peak
current limit. This action becomes the switch current limit
specifi cation. The maximum available output power is then
determined by the switch current limit.
A potential controllability problem could occur under
short-circuit conditions. If the power supply output is
short circuited, the feedback amplifi er responds to the
low output voltage by raising the control voltage, V
C
,
to its peak current limit value. Ideally, the output switch
would be turned on, and then turned off as its current
exceeded the value indicated by V
C
. However, there is fi nite
response time involved in both the current comparator and
turn-off of the output switch. These result in a minimum
on-time, t
ON(MIN)
. When combined with the large ratio of
V
IN
to (V
F
+ I • R), the diode forward voltage plus inductor
I • R voltage drop, the potential exists for a loss of control.
Expressed mathematically the requirement to maintain
control is:
ft
VIR
V
ON
F
IN
•
•
≤
+
where:
f = Switching frequency
t
ON
= Switch minimum on-time
V
F
= Diode forward voltage
V
IN
= Input voltage
I • R = Inductor I • R voltage drop
If this condition is not observed, the current will not be
limited at I
PK
, but will cycle-by-cycle ratchet up to some
higher value. Using the nominal LT1766 clock frequency
of 200KHz, a V
IN
of 40V and a (V
F
+ I • R) of say 0.7V, the
maximum t
ON
to maintain control would be approximately
90ns, an unacceptably short time.
The solution to this dilemma is to slow down the oscil-
lator when the FB pin voltage is abnormally low thereby
indicating some sort of short-circuit condition. Oscillator
frequency is unaffected until FB voltage drops to about
2/3 of its normal value. Below this point the oscillator
<
+()(––)
()( )()()
VVVVV
VfL
OUT F IN OUT F
IN
2
=
=
()( )
+
I
IfLV
VVVVV
P
PIN
OUT F IN OUT F
2
2
2
2
()( )
()( )( )
()(––)
I
LP-P
=
+
(.)•(•)( )()
(.)(––.)
–
1 5 200 10 10 15
25063155063
235
APPLICATIONS INFORMATION
LT1766/LT1766-5
15
1766fc
capacitors fail during very high
turn-on
surges, which
do not occur at the output of regulators. High
discharge
surges, such as when the regulator output is dead shorted,
do not harm the capacitors.
Unlike the input capacitor, RMS ripple current in the output
capacitor is normally low enough that ripple current rating
is not an issue. The current waveform is triangular with
a typical value of 125mA
RMS
. The formula to calculate
this is:
Output capacitor ripple current (RMS):
I
VVV
LfV
RIPPLE RMS
OUT IN OUT
IN
()
=
()
−
()
()()( )
029.
Ceramic Capacitors
Higher value, lower cost ceramic capacitors are now
becoming available. They are generally chosen for their
good high frequency operation, small size and very low
ESR (effective series resistance). Their low ESR reduces
output ripple voltage but also removes a useful zero in the
loop frequency response, common to tantalum capaci-
tors. To compensate for this, a resistor R
C
can be placed
in series with the V
C
compensation capacitor, C
C
. Care
must be taken however, since this resistor sets the high
frequency gain of the error amplifi er, including the gain at
the switching frequency. If the gain of the error amplifi er
is high enough at the switching frequency, output ripple
voltage (although smaller for a ceramic output capacitor)
may still affect the proper operation of the regulator. A
fi lter capacitor, C
F
, in parallel with the R
C
/C
C
network is
suggested to control possible ripple at the V
C
pin. An All
Ceramic solution is possible for the LT1766 by choos-
ing the correct compensation components for the given
application.
Example: For V
IN
= 8V to 40V, V
OUT
= 3.3V at 1A, the
LT1766 can be stabilized, provide good transient response
and maintain very low output ripple voltage using the
following component values: (refer to the fi rst page of
this data sheet for component references) C3 = 2.2μF,
R
C
= 4.7k, C
C
= 15nF, C
F
= 220pF and C1 = 47μF. See
Application Note 19 for further detail on techniques for
proper loop compensation.
frequency decreases roughly linearly down to a limit
of about 40kHz. This lower oscillator frequency during
short-circuit conditions can then maintain control with
the effective minimum on time.
It is recommended that for [V
IN
/(V
OUT
+ V
F
)] ratios > 10,
a soft-start circuit should be used to control the output
capacitor charge rate during start-up or during recovery
from an output short circuit, thereby adding additional
control over peak inductor current. See Buck Converter
with Adjustable Soft-Start later in this data sheet.
OUTPUT CAPACITOR
The output capacitor is normally chosen by its effective
series resistance (ESR), because this is what determines
output ripple voltage. To get low ESR takes
volume
, so
physically smaller capacitors have high ESR. The ESR
range for typical LT1766 applications is 0.05Ω to 0.2Ω.
A typical output capacitor is an AVX type TPS, 100μF at
10V, with a guaranteed ESR less than 0.1Ω. This is a “D”
size surface mount solid tantalum capacitor. TPS capaci-
tors are specially constructed and tested for low ESR, so
they give the lowest ESR for a given volume. The value
in microfarads is not particularly critical, and values from
22μF to greater than 500μF work well, but you cannot
cheat mother nature on ESR. If you fi nd a tiny 22μF solid
tantalum capacitor, it will have high ESR, and output ripple
voltage will be terrible. Table 2 shows some typical solid
tantalum surface mount capacitors.
Table 3. Surface Mount Solid Tantalum Capacitor ESR
and Ripple Current
E Case Size ESR (MAX, Ω ) RIPPLE CURRENT (A)
AVX TPS, Sprague 593D 0.1 to 0.3 0.7 to 1.1
D Case Size
AVX TPS, Sprague 593D 0.1 to 0.3 0.7 to 1.1
C Case Size
AVX TPS 0.2 (typ) 0.5 (typ)
Many engineers have heard that solid tantalum capacitors
are prone to failure if they undergo high surge currents. This
is historically true, and type TPS capacitors are specially
tested for surge capability, but surge ruggedness is not
a critical issue with the
output
capacitor. Solid tantalum
APPLICATIONS INFORMATION
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LT1766IGN-5#TRPBF

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Buy LT1766IGN-5#TRPBF
Manufacturer:
Analog Devices / Linear Technology
Description:
Switching Voltage Regulators 1.5A 200kHz High Voltage Step-down Regulator
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