IXMS150PSI Datasheet IXMS150PSI | P4-P6

I - 38

IXMS 150

Functional Description

Introduction

The IXMS150 is designed with mono-

lithic CMOS technology. The IC is

primarily intended for use with two-

phase step motors in the microstepping

mode but may also be used for control

of two DC motors, audio amplifiers, or

any application requiring two synchro-

nized PWMs. The IXMS150 simulta-

neously controls the currents in each of

two separate H-bridges. This device

utilizes

both analog and digital functions.

The IC has five fundamental sections:

(1) oscillator and feedforward circuitry,

(2) analog section for control of the

motor currents, (3) a protection network

to protect the H-bridges and the motor

from abnormal conditions, (4) the digital

PWM logic for the control signals, and

(5) the power supply section which

includes a negative bias generator.

Oscillator

The IXMS150 contains an internal

oscillator which is controlled by

adjusting the values of R

and C

These two components determine the

switching frequency, amount of dead

time, and the minimum pulse width at

output pins 9, 10, 15 and 16. The

minimum and maximum values of R

and C

are given in the Electrical

Characteristics.

The oscillator also sets the frequency of

the charge pump circuit in the internal

negative bias generator (V

). At lower

frequencies (<40 kHz) the value of

CPUMP must be increased to assure

proper operation.

Feedforward Compansation

In all fixed frequency PWM control

systems open loop gain, motor current

slew rate, and motor current ripple are

proportional to the motor supply

voltage. Gain variations due to supply

voltage changes complicate the design

of such systems and restrict their band-

width to the minimum worst case

condition. For this reason, an advanced

adaptive compensation scheme is built-

in using a feedforward technique. This

feature has been designed such that

open loop gain is inversely proportional

to the voltage applied to the FFWD pin,

normally a fraction of the motor supply.

As a result, open loop gain can be

made independent of the high voltage

supply and system bandwidth can be

maximized.

Analog Section

The analog section of each channel of

the IXMS150 consists of a signal

processor and an error amplifier. The

signal processor is required since the

voltage developed across the sense

resistor often contains transients asso-

ciated with the switching characteristics

of the power devices. These transients

need to be properly filtered for the

system to operate with the desired

degree of precision. Because of this,

the IXMS150 uses proprietary analog

and digital signal processing techni-

ques that sense the true average phase

currents. Since this requires only one

sense resistor per H-bridge it avoids

mismatches in charge/discharge

currents associated with two sense

resistor per H-bridge topologies.

The instantaneous difference between

the motor current and the control input

is integrated via the E/A amp and fed to

the PWM comparator to generate the

appropriate signals for the H-bridges.

External compensation of the input and

sense signals is provided for via the

comp1, comp2 and comp3 pins.

Protection Circuitry

The IC has a two-level Over/Excess

Current protection circuit. Maximum

current is represented as 0.625 V at the

SENSE input. If the SENSE voltage

exceeds 0.9 volts for more than one

microsecond, the switching outputs

(VOUT) and OUTDIS will be forced low.

This represents a current that is 40 %

beyond full scale. If the SENSE voltage

exceeds 3.6 V, these outputs will be

forced low immediately. This repre-

sents a current that is 500% beyond full

scale. The time delay on the lower level

of overcurrent avoids erroneous

shutdowns as a result of noise spikes

that are coupled from the motors H-

bridges. Note that the threshold

voltages cited here assume a supply of

+12 V.

Undervoltage Lockout

A third protection mechanism is the

Under-Voltage Lockout. It assures

proper behavior on power-up and

power-down and avoids high power

dissipation in the H-bridge due to

insufficient gate voltage. It uses a zener

for reference and has a trip point set at

8 V. It will also check to make sure

there is sufficient negative bias to

insure proper operation. This is typically

-1.6 V. OUTDIS will be held low by the

UV Lockout circuit until V

and V

reach these values.

Output Disable Feature

To enable external over-temperature

protection, the output disable pin

(OUTDIS) is available on the IXMS150.

When pulled low this disables the

output by forcing all output pins low.

The same output disable input pin is

also used as a status output. When it is

pulled low by the internal circuitry it

indicates an error condition such as

undervoltage (V

), insufficient negative

bias voltage (V

) or over/excess

current. This can be used as a status

indicator in smart systems.

PWM Section

The PWM comparator generates two

complementary signals based on the

output of the error amplifier. Dead-time

is then added which is adjusted by the

selection of the external oscillator

capacitor. There is also a minimum duty

cycle clamp circuit that allows the use

of an AC coupled H-bridge.

Supply Section

The main power supply (V

) is applied

to pin 24. This is typically +12 V. Internal

bias circuitry presents a V

/2 reference

voltage at pin 8, BYPASS. A 0.1 µF

capacitor should be connected from pin

8 to analog ground for noise immunity.

Negative Bias Generator

The IXMS150 samples both positive

and negative voltages at the motor

sense feedback resistor. In addition,

since errors in the input current around

zero are a major contributor to micro-

step positioning error, the input control

range is bipolar and specified as ±2 V

full scale. For these reasons it is desi-

rable to have both positive and nega-

tive power supplies. In order to enable

single 12 V supply operation, a negative

voltage generator and regulator are

built into the IC. This is a charge pump

circuit whose frequency is that of the

onboard oscillator. It utilizes an external

pair of capacitors and diodes to gene-

rate a negative bias equal to -V

/5 or

approximately -2.4 V for V

= 12 V.

I - 39

Application Information

Introduction

The advantages of step motors are well

known. They may be operated in an

open loop fashion, the accuracy of

which is mostly dependent on the

mechanical accuracy of the motor. They

move in quantized increments (steps)

which lends them easily to digitally

controlled motion systems. In addition,

their drive signals are square wave in

nature and are therefore easily gene-

rated with relatively high efficiency due

to their ON/OFF characteristics.

But step motors are not free of prob-

lems. Their large pulse drive wave-

forms create mechanical forces which

excite and aggravate the mechanical

resonances in the system. These are

load dependent and difficult to control

since step motors have very little

damping of their own. At resonance a

step motor system is likely to lose

synchronization and therefore skip or

gain a

step. Being an open loop system,

this would imply loss of position infor-

mation and would be unacceptable. A

common method of solving this problem

is to avoid the band of resonance

frequencies altogether, but this might

put severe limitations on system

performance. Steppers have 200 steps

per revolution or 1.8 degrees per step.

The highest resolution commercially

available steppers have 400 steps per

revolution or 0.9 degrees per step.

Microstepping Mode

One way to circumvent the problems

associated with step motors while still

retaining their open loop advantages is

to use them in the microstepping mode.

In this mode each of the steps is subdi-

vided

into smaller steps or microsteps".

Applying currents to both phases of the

motor creates a torque phaser which is

proportional to the vector sum of both

currents. When the phasor completes

one turn (360 electrical degrees), the

motor moves exactly four full steps or

one torque cycle. Similarly, when that

phasor moves 22.5 electrical degrees

the motor will move (22.5/90)  100 =

25 % of a full step. Thus the position of

the motor is determined by the angle of

the torque phasor. When used with an

appropriate motor a positioning accu-

racy of 2 % of a full step can be achie-

ved, equaling 0.036 degrees for a 200

full steps per revolution motor. In this

manner the motor can be positioned to

any arbitrary angle. A common way to

control the angle of the torque phasor is

by applying to the motors phases two

periodic waveforms shifted by 90

electrical degrees.

Let the phase current equations be:

= I

 cos θe (1)

= I

 sin θe (2)

Note that θe is the electrical position.

The resulting torque generated by the

corresponding phases would then be:

= K

 i

= K

 I

 cos θe (3)

= K

 i

= K

 I

 sin θe (4)

where K

is the torque constant of the

motor. Substituting Eqs. (1), (2) into (3),

(4) and doing vector summation the

resulting total generated torque mea-

sured on the motor shaft is given by:

= K

 I

(5)

Note that in this case we have zero

torque ripple.

Using this technique one can theore-

tically achieve infinite resolution with

any step motor. Since the drive current

waveforms are sinusoidal instead of

square, the step to step oscillations are

eliminated and the associated velocity

ripple. This greatly improves perfor-

mance at low rotational speeds and

helps avoid resonance problems. In an

actual application, the extent to which

these things are true depends on how

the two sinusoidal reference waveforms

are generated.

Seemingly we have lost the quantized

motion feature of a stepper when used

in this mode. This can be regained by

defining the term microsteps per step.

Each full step is subdivided into micro-

steps by applying to the motors phases

those intermediate current levels for

which their vector sum tracks the circle

of Fig. 2 and divides the full step (90

electrical degrees) into the require

number of microsteps. An example of

the required phase currents for full step

and four microstep per step operation

are shown in Fig. 1 and 2 respectively.

Phase Current Matching

Requirements

Assuming microstepping is being used

for resolution improvement and not as a

resonance avoidance technique, a step

motor can be selected knowing the

torque needed, its specified step

IXMS 150

Fig. 1 Full Step Drive Waveforms

accuracy, and the required resolution or

the number of microsteps per step.

Next, one must determine the accuracy

required of the phase currents to main-

tain the accuracy of the complete

system. Equations (1) - (4) clearly

indicate that errors in the absolute

value or phase of the phase currents

will impact positioning accuracy.

Another observation is that by keeping

the ratio of the phase currents i

constant, errors in their value will result

Fig. 2 Four Microstep per Step Drive

Waveforms

I - 40

IXMS 150

the H-bridge that must be properly

filtered if the system is to operate with

the desired degree of precision.

This presents a significant engineering

challenge that has been solved by

IXYSs design team. Using proprietary

analog and digital signal processing

techniques, IXYS has developed a

control system that measures the true

average phase currents. Requiring only

one sense resistor per H-bridge, this

technique avoids errors due to mis-

matches in charge/discharge currents

associated with using one sense resi-

stor on each leg of the H-bridge. This

improves system performance as well

as minimizing component count. The

sense resistor for each H-bridge should

be selected based on the required peak

motor current:

= 0.625 V/I

mpk

(9)

The voltage developed across this

resistor is then applied to the corres-

ponding sense input for each H-bridge.

Negative bias Generator

One of todays cost cutting trends is to

minimize the number of power supplies,

implying single supply operation for the

control section. Yet the current feed-

back and reference inputs are bipolar

signals. Level shifting has been used

for the reference input in the past, but

that can not be easily done for the

feedback signal without impacting

accuracy or efficiency. In practice one

finds that in order to generate true zero

voltage having low impedance drive

there must be a negative power supply.

Otherwise there will be a tradeoff

sacrificing accuracy for simpler system

design.

For these reasons the approach selec-

ted by IXYS was different. Taking

advantage of our CMOS design, we

opted to build into the chip a negative

bias generator. This does put stringent

demands on noise coupling but results

in the most flexible system having the

highest possible accuracy. The built in

charge pump circuit requires two capa-

citors and two diodes to be added

externally. The recommended compo-

nent values for an oscillator frequency

of 100 kHz are given below.

C1 = 0.047 µF

C2 = 100 µF

D1 = D2 = 1N4148

Note: V

= -(V

/5)

in torque value errors but no positioning

errors. The question is, what is the

upper bound on the current errors in

order to keep the position error within

some given angle ∆θ.

Referring to Fig.

3, assume the required

currents i

, i

are given by Equations

(1), (2) respectively such that their

vector sum points to position P. Let the

phase currents vary by a small amount

such that their vector sum lies within a

circle centered at point P and having

the radius i, as indicated in Fig. 3.

Fig. 3 Effect of Current Errors on

Position

If follows that the worst case position

error occurs for the cases where the

vector sum is tangent to the circle such

as point P

, at which:

tan (∆ θ) = ∆ i/l

(6)

For instance, to keep position error to

less than 1% of a full step, the electrical

angular error would be:

∆ θ = 0.01  90° = 0.9° (7)

This is assuming there are 90 electrical

degrees for a full step. Therefore total

current error must be:

i/I

= tan (∆θ) = 0.016 or 1.6 %) (8)

Thus the current error must be kept to

less than 1.6 % of full scale or peak

current at each phase for 1

% maximum

position error. This upper bound on

error includes all error sources such as

zero offset errors and full scale matching

errors. Another interesting observation

is that in the vicinity of a full step (i.e.,

θe = 0), the phase having the bigger

impact on position error is the one

carrying the smaller current through it.

This has a strong impact on input

waveform generation.

Input waveform generation

It has been shown that the two input

signals, VIN

and VIN

, are sinusoidal

and 90° out of phase. This may be

accomplished by using two look-up

Fig. 4 Simple Reference Waveform

Generator

tables stored in ROM and two DACs

per Fig. 4. An up/down counter may be

used to generate the appropriate

address locations for the ROMs and the

data outputs used to control the DACs.

The user then need only supply up or

down pulses to the counter to control

the IXMS150 and hence the motor.

In higher performance systems a

microprocessor may be used in place

of the counter and the ROMs. The

micro can perform the look-up function

and calculate the appropriate system

responses, velocity profiles, etc.

necessary for total system operation.

An example of this configuration is

shown in Fig. 5.

Current Sensing Considerations

Most commercially available monolithic

PWM controllers monitor and control

the peak of the phase current by com-

paring the voltage across the sense

resistor with a ramp voltage. This

approach assumes that the ripple

current is fixed in amplitude. Results

shown later clearly indicate the varia-

tion of the ripple current with frequency.

But even in fixed frequency systems

the ripple current is directly proportional

to the motor supply voltage and to the

back EMF voltage of the motor. Ripple

current is not insignificant compared to

the full scale current and therefore

cannot be neglected in a precision

system. In addition, there are transients

associated with the turn on and turn off

characteristics of the power devices in

Fig. 5 Microprocessor Based

Referenced Waveform Generator

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