NCP1124AP65G Datasheet NCP1129AP100G, NCP1126BP65G, NCP1124BP100G | P13-P15

NCP1124, NCP1126, NCP1129

www.onsemi.com

APPLICATION INFORMATION

Introduction

The NCP112x family integrates a high−performance

current−mode controller with a 650 V MOSFET, which

considerably simplifies the design of a compact and reliable

switch mode power supply (SMPS). This component

represents the ideal candidate where low part−count and cost

effectiveness are the key parameters. The NCP112x brings

most necessary functions needed in today’s modern power

supply designs, with several enhancements such as V

OVP, adjustable slope compensation, frequency jittering,

frequency foldback, skip cycle, etc.

• Current−mode operation with adjustable internal

ramp compensation: Sub−harmonic oscillations in

peak current mode control can be eliminated by the

adjustable internal ramp compensation when the duty

ratio is larger than 0.5.

• Frequency foldback capability: When the load

current drops, the controller responds by reducing the

primary peak current. When the peak current reaches

the skip peak current level, the NCP112x enter skip

operation to reduce the power consumption.

• Internal soft−start: a soft−start precludes the main

power switch from being stressed upon start−up. In this

switcher, the soft−start is internally fixed to 4 ms.

Soft−start is activated when a new startup sequence

occurs or during an auto−recovery hiccup.

• Latched OVP on V

: When the V

exceeds 28 V

typical, the drive signal is disabled and the part latches

off. When the user cycles the V

down, the circuit is

reset and the part enters a new start up sequence.

• Short−circuit protection: short−circuit and especially

over−load protections are difficult to implement when a

strong leakage inductance between the auxiliary and the

power windings affects the transformer (the aux

winding level does not properly collapse in presence of

an output short). Every time the internal 0.8 V

maximum peak current limit is activated, an error flag

is asserted and an internal timer starts. When the fault is

validated, the switcher will either be latched or enter

the auto−recovery mode. As soon as the fault

disappears, the SMPS resumes operation.

• EMI jittering: an internal low−frequency 240 Hz

modulation signal varies the pace at which the

oscillator frequency is modulated. This helps spread out

the energy in a conducted noise analysis. To improve

the EMI signature at low power levels, the jittering will

not be disabled in frequency foldback mode (light load

conditions).

Start−up Sequence

The NCP112x need an external startup circuit to provide

the initial energy to the switcher. As is shown in Figure 39,

the startup circuit consists of R

start

and V

capacitor C

connected to the main input, i.e. half−wave connection. The

auxiliary winding will take over the RC circuit after the

output voltage is built up.

Auxiliary

winding

Main

Input

Figure 39. Startup Circuit for NCP112x

(half−wave connection)

bulk

start

The startup process can be well explained by Figure 40. At

power on, when the V

capacitor is fully discharged, the

switcher current consumption is zero and does not deliver

any driving pulses. The V

capacitor C

is going to be

charged by the main input via R

start

. As V

increases, the

switcher consumed current remains below a guaranteed

limit until the voltage on the capacitor reaches V

CC(on)

, at

which point the switcher starts to deliver pulses to the power

MOSFET. The switcher current consumption suddenly

increases, and the capacitor depletes since it is the only

energy reservoir. Its voltage falls until the auxiliary winding

takes over and supply the V

pin.

Drive

time

margin

Figure 40. Startup Process for NCP112x

CC(off)

CC(on

)

: 5−20 ms

The start−up current of the switcher is extremely low,

below 15 mA. The start−up resistor can be connected to the

bulk capacitor or directly the mains input voltage for further

power dissipation reduction. The switcher begins switching

when V

reaches V

CC(on)

, typically 17 V for NCP1126/9.

From Figure 41, it can be seen that the startup resistor R

start

and V

capacitor are about to be determined.

NCP1124, NCP1126, NCP1129

www.onsemi.com

Capacitor

The supply capacitor, C

, provides power to the switcher

during power up. The capacitor must be large enough such

that a V

voltage greater than V

CC(off)

is maintained while

the auxiliary supply voltage is building up. Otherwise, V

will collapse and the switcher will turn off. Assuming this

time t

is equal to 10 ms, Equation 1 is used to calculate the

required V

capacitor.

CC(on)

* V

CC(off)

(eq. 1)

Startup Resistor R

start

In order to determine the startup resistor, the V

capacitor charging current is calculated first to ensure that

the charging time for the V

capacitor from 0 V to its

operating voltage meets the startup time requirement.

Equation 2 gives the first constraints for the R

start

selection.

charge

CC(on)

startup

(eq. 2)

For NCP1126/9, during startup process, from 0 to t

, the

current that flow inside the switcher is I

CC1

, therefore the

total charging current from the main input is going to be I

= I

charge

+ I

CC1

. Consider the half−wave connection start−up

network to the mains as is shown in Figure 41, the average

current flowing into this start−up resistor will be the smallest

when V

reaches the V

CC(on)

of the switcher:

c,min

ac,rms

* V

CC(on)

start−up

(eq. 3)

which gives the minimum value for the R

startup

start−up

ac,rms

* V

CC(on)

c,min

(eq. 4)

Note that this calculation is purely theoretical, considering

a constant charging current. In reality, the take over time can

be shorter (or longer!) and it can lead to a reduction of the

capacitor. This brings a decrease in the charging current

and an increase of the start−up resistor, for the benefit of

standby power. The dissipated power at high line amounts

to:

diss

ac,peak

start

(eq. 5)

The above derivation is based on the case when the power

supply is not at light load. V

capacitor selection should

ensure that does not disappear in no−load conditions. In light

load condition, the skip−cycle can be so deep that refreshing

pulses are likely to be widely spaced, inducing a large ripple

on the V

capacitor. If this ripple is too large, chances exist

to hit the V

CC(off)

and reset the switcher into a new start−up

sequence. A solution is to grow this capacitor but it will

obviously be detrimental to the start−up time. The option

offered in Figure 41 elegantly solves this potential issue by

adding an extra capacitor C

CC,aux

on the auxiliary winding.

However, this component is separated from the V

pin by

a simple diode. You therefore have the ability to grow this

capacitor as you need to ensure the self−supply of the

switcher without affecting the start−up time and standby

power.

Auxiliary

winding

Main

Input

Vcc

Figure 41. Startup Circuit for NCP112x (half−wave

connection), Considering Light Load Condition

start

CC,aux

bulk

Frequency Foldback

The reduction of no−load standby power associated with

the need for improving the efficiency, requires a change in

the traditional type of fixed−frequency operation. NCP112x

implement a switching frequency foldback function when

the feedback voltage is below V

FB(fold)

. At this point, the

oscillator turns into a Voltage−Controlled Oscillator and

reduces its switching frequency. The peak current setpoint

follows the feedback pin until its level reaches V

FB(freeze)

Below this value, the peak current freezes to V

FB(freeze)

/ 4.

The operating frequency is down to f

trans

when the feedback

voltage reaches V

FB(fold,end)

. Below this point, if the output

power continues to decrease, the part enters skip mode for

the best noise−free performance in no−load conditions.

Figure 6 depicts the adopted scheme for the part.

Over−voltage Protection

The latched−state of the NCP112x is maintained via an

internal thyristor (SCR). When the voltage on pin 1 exceeds

the latch voltage for four consecutive clock cycles, the SCR

is fired and immediately stops the output pulses. The same

SCR is fired when an OVP is sensed on the V

pin. When

this happens, all pulses are stopped and V

is discharged

to a fix level of 7 V typically: the circuit is latched and the

converter no longer delivers pulses. To maintain the

latched−state, a permanent current must be injected in the

part. If too low of a current, the part de−latches and the

converter resumes operation. This current is characterized to

32 mA as a minimum but we recommend including a design

margin and select a value around 60 mA. The test is to latch

the part and reduce the input voltage until it de−latches. If

you de−latch at V

= 70 V

rms

for a minimum voltage of

85 V

rms

, you are fine.

NCP1124, NCP1126, NCP1129

www.onsemi.com

min

max

trans

min

max

3.2 V

Frequency

Figure 42. Frequency Foldback Architecture

ILIM

CS(fold)

CS(freeze)

FB(freeze)

FB(fold)

FB(fold,end)

OSC

If it precociously recovers, you will have to increase the

start−up current, unfortunately to the detriment of standby

power.

The most sensitive configuration is actually that of the

half−wave connection proposed in Figure 39. As the current

disappears 5 ms for a 10 ms period (50 Hz input source), the

latch can potentially open at low line. If you really reduce the

start−up current for a low standby power design, you must

ensure enough current in the SCR in case of a faulty event.

An alternate connection to the above is shown in Figure 43:

Figure 43. The Full−wave Connection Ensures Latch

Current Continuity as Well as a X2−Discharge Path

In this case, the current is no longer made of 5 ms “holes”

and the part can be maintained at a low input voltage.

Experiments show that these 2−MW resistor help to maintain

the latch down to less than 50 V rms, giving an excellent

design margin. Standby power with this approach was also

improved compared to Figure 39 solution. Please note that

these resistors also ensure the discharge of the X2−capacitor

up to a 0.47 mF type.

The de−latch of the SCR occurs when a) the injected

current in the V

pin falls below the minimum stated in the

data−sheet (32 mA at room temp) or when the part senses a

brown−out recovery.

Auto−Recovery Short−Circuit Protection

In case of output short−circuit or severe overload

situation, an internal error flag is raised and starts a

countdown timer. If the flag is asserted longer than t

OVLD

the driving pulses are stopped and V

falls down as the

auxiliary pulses are missing. When it hits V

CC(off)

, the

switcher consumption is down to a few mA and the V

slowly builds up again by the startup network R

start

, C

When V

reaches V

CC(on)

, the switcher purposely ignores

the re−start and waits for another V

cycle: this is the

so−called double hiccup. Illustration of such principle

appears in Figure 13. Please note that soft−start is activated

upon re−start attempt.

drive

time

Figure 44. Auto−Recovery Double Hiccup Sequence

CC(off)

CC(on)

P1-P3 P4-P6 P7-P9 P10-P12 P13-P15 P16-P18

NCP1124AP65G

Mfr. #:

Buy NCP1124AP65G

Manufacturer:

ON Semiconductor

Description:

AC/DC Converters HV SWITCHER FOR OFFLINE P

Lifecycle:

New from this manufacturer.

Delivery:

DHL FedEx Ups TNT EMS

Payment:

T/T Paypal Visa MoneyGram Western Union

NCP1124AP65G

NCP1124AP65G

Products related to this Datasheet