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LT8302ES8E#PBF

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LT8302
13
8302fc
For more information www.linear.com/LT8302
applicaTions inForMaTion
Output Power
A flyback converter has a complicated relationship between
the input and output currents compared to a buck or a
boost converter. A boost converter has a relatively constant
maximum input current regardless of input voltage and a
buck converter has a relatively constant maximum output
current regardless of input voltage. This is due to the
continuous non-switching behavior of the two currents. A
flyback converter has both discontinuous input and output
currents which make it similar to a nonisolated buck-boost
converter. The duty cycle will affect the input and output
currents, making it hard to predict output power. In ad
-
dition, the winding ratio can be changed to multiply the
output current at the expense of a higher switch voltage.
The graphs in Figures 1 to 4 show the typical maximum
output power possible for the output voltages 3.3V, 5V,
12V, and 24V. The maximum output power curve is the
calculated output power if the switch voltage is 50V dur
-
ing the
switch-off time. 15V of margin is left for leakage
inductance voltage spike. To
achieve this power level at
a given input, a winding ratio value must be calculated
to stress the switch to
50V, resulting in some odd ratio
values. The curves below the maximum output power
curve are examples of common winding ratio values and
the amount of output power at given input voltages.
One design example would be a 5V output converter with
a minimum input voltage of 8V and a maximum input volt
-
age of 32V. A three-to-one winding ratio fits this design
example perfectly and outputs equal to 15.3W at 32V but
lowers to 7.7W at 8V.
Figure 1. Output Power for 3.3V Output
Figure 2. Output Power for 5V Output
Figure 3. Output Power for 12V Output
Figure 4. Output Power for 24V Output
INPUT VOLTAGE (V)
0
OUTPUT POWER (W)
10
15
40
8302 F02
5
0
10
20
30
MAXIMUM
OUTPUT POWER
20
N = 3:1
N = 1:1
N = 4:1
N = 2:1
INPUT VOLTAGE (V)
0
OUTPUT POWER (W)
10
15
40
8302 F03
5
0
10
20
30
N = 1:1
MAXIMUM
OUTPUT POWER
20
N = 3:2
N = 1:2
N = 2:1
INPUT VOLTAGE (V)
0
OUTPUT POWER (W)
10
15
40
8302 F04
5
0
10
20
30
N = 1:2
MAXIMUM
OUTPUT POWER
20
N = 2:3
N = 1:3
N = 1:1
INPUT VOLTAGE (V)
0
OUTPUT POWER (W)
10
15
40
8302 F01
5
0
10
20
30
N = 6:1
MAXIMUM
OUTPUT POWER
20
N = 4:1
N = 2:1
N = 3:1
LT8302
14
8302fc
For more information www.linear.com/LT8302
applicaTions inForMaTion
The equations below calculate output power:
P
OUT
= η • V
IN
• D • I
SW(MAX)
• 0.5
η = Efficiency = ~85%
D= Duty Cycle =
V
OUT
+ V
F
( )
•N
PS
V
OUT
+ V
F
( )
•N
PS
+ V
IN
I
SW(MAX)
= Maximum switch current limit = 3.6A (MIN)
Primary Inductance Requirement
The LT8302 obtains output voltage information from the
reflected output voltage on the SW pin. The conduction
of secondary current reflects the output voltage on the
primary SW pin. The sample-and-hold error amplifier needs
a minimum 350ns to settle and sample the reflected output
voltage. In order to ensure proper sampling, the second
-
ary winding
needs to conduct current for a minimum of
350ns. The following
equation gives the minimum value
for primary-side magnetizing inductance:
L
PRI
≥
t
OFF(MIN)
•N
PS
• V
OUT
+ V
F
( )
I
SW(MIN)
t
OFF(MIN)
= Minimum switch-off time = 350ns (TYP)
I
SW(MIN)
= Minimum switch current limit = 0.87A (TYP)
In addition to the primary inductance requirement for
the minimum switch-off time, the LT8302 has minimum
switch-on time that prevents the chip from turning on
the power switch shorter than approximately 160ns. This
minimum switch-on time is mainly for leading-edge blank
-
ing the initial switch turn-on current spike. If the inductor
current exceeds the desired current limit during that time,
oscillation may occur at the output as the current control
loop will lose its ability to regulate. Therefore, the following
equation relating to maximum input voltage must also be
followed in selecting primary-side magnetizing inductance:
L
PRI
≥
t
ON(MIN)
• V
IN(MAX)
I
SW(MIN)
t
ON(MIN)
= Minimum switch-on time = 160ns (TYP)
In general, choose a transformer with its primary mag-
netizing inductance
about 40%
to 60% larger than the
minimum values calculated above. A transformer with
much larger inductance will have a bigger physical size
and may cause instability at light load.
Selecting a Transformer
Transformer specification and design is perhaps the most
critical part of successfully applying the LT8302. In addition
to the usual list of guidelines dealing with high frequency
isolated power supply transformer design, the following
information should be carefully considered.
Linear Technology has worked with several leading mag
-
netic component
manufacturers to
produce pre-designed
flyback transformers for use with the LT8302. Table 1
shows the details of these transformers.
Table 1. Predesigned Transformers–Typical Specifications
TRANSFORMER
PART NUMBER
DIMENSIONS
(W × L × H) (mm)
L
PRI
(µH)
L
LKG
(µH) N
P
:N
S
R
PRI
(mΩ)
R
SEC
(mΩ) VENDOR
TARGET APPLICATION
V
IN
(V) V
OUT
(V) I
OUT
(A)
750311625 17.75 × 13.46 × 12.70 9 0.35 4:1 43 6 Würth Elektronik 8 to 32 3.3 2.1
750311564 17.75 × 13.46 × 12.70 9 0.12 3:1 36 7 Würth Elektronik 8 to 32 5 1.5
750313441 15.24 × 13.34 x 11.43 9 0.6 2:1 75 18 Würth Elektronik 8 to 32 5 1.3
750311624 17.75 × 13.46 × 12.70 9 0.18 3:2 34 21 Würth Elektronik 8 to 32 8 0.9
750313443 15.24 × 13.34 × 11.43 9 0.3 1:1:1 85 100 Würth Elektronik 8 to 36 ±12 0.3
750313445 15.24 × 13.34 × 11.43 9 0.25 1:2 85 190 Würth Elektronik 8 to 36 24 0.3
750313457 15.24 × 13.34 × 11.43 9 0.25 1:4 85 770 Würth Elektronik 8 to 36 48 0.15
750313460 15.24 × 13.34 × 11.43 12 0.7 4:1 85 11 Würth Elektronik 4 to 18 5 0.9
750311342 15.24 × 13.34 × 11.43 15 0.44 2:1 85 22 Würth Elektronik 4 to 18 12 0.4
750313439 15.24 × 13.34 × 11.43 12 0.6 2:1 115 28 Würth Elektronik 18 to 42 3.3 2.1
750313442 15.24 × 13.34 × 11.43 12 0.75 3:2 150 53 Würth Elektronik 18 to 42 5 1.6
LT8302
15
8302fc
For more information www.linear.com/LT8302
applicaTions inForMaTion
Turns Ratio
Note that when choosing an R
FB
/R
REF
resistor ratio to set
output voltage, the user has relative freedom in selecting
a transformer turns ratio to suit a given application. In
contrast, the use of simple ratios of small integers, e.g.,
3:1, 2:1, 1:1, etc., provides more freedom in settling total
turns and mutual inductance.
Typically, choose the transformer turns ratio to maximize
available output power. For low output voltages (3.3V
or 5V), a N:1 turns ratio can be used with multiple pri
-
mary windings relative to the secondary to maximize the
transformer’s current gain (and output power). However,
remember that the SW pin sees a voltage that is equal
to the maximum input supply voltage plus the output
voltage multiplied by the turns ratio. In addition, leakage
inductance will cause a voltage spike (V
LEAKAGE
) on top of
this reflected voltage. This total quantity needs to remain
below the 65V absolute maximum rating of the SW pin to
prevent breakdown of the internal power switch. Together
these conditions place an upper limit on the turns ratio,
N
PS
, for a given application. Choose a turns ratio low
enough to ensure
N
PS
<
65V – V
IN(MAX)
– V
LEAKAGE
V
OUT
+ V
F
For larger N:1 values, choose a transformer with a larger
physical size to deliver additional current. In addition,
choose a large enough inductance value to ensure that
the switch-off time is long enough to accurately sample
the output voltage.
For lower output power levels, choose a 1:1 or 1:N trans
-
former for the absolute smallest transformer size. A 1:N
transformer will minimize the magnetizing inductance
(and minimize size), but will also limit the available output
power. A higher 1:N turns ratio makes it possible to have
very high output voltages without exceeding the breakdown
voltage of the internal power switch.
The turns ratio is an important element in the isolated
feedback scheme, and directly affects the output voltage
accuracy. Make sure the transformer manufacturer speci
-
fies turns ratio accuracy within ±1%.
Saturation
Current
The current in
the transformer windings should not exceed
its rated saturation current. Energy injected once the core is
saturated will not be transferred to the secondary and will
instead be dissipated in the core. When designing custom
transformers to be used with the LT8302, the saturation
current should always be specified by the transformer
manufacturers.
Winding Resistance
Resistance in either the primary or secondary windings
will reduce
overall power efficiency. Good
output voltage
regulation will be maintained independent of winding re-
sistance due
to the boundary/discontinuous conduction
mode operation of the LT8302.
Leakage Inductance and Snubbers
Transformer leakage
inductance on either the primary or
secondary causes a voltage spike to appear on the primary
after the power switch turns off. This spike is increasingly
prominent at higher load currents where more stored en
-
ergy must be dissipated. It is very important to minimize
transformer leakage inductance.
When designing an
application, adequate margin should
be kept for the worst-case leakage voltage spikes even
under overload conditions. In most cases shown in Fig
-
ure5, the
reflected output voltage on the primary plus V
IN
should be kept below 50V. This leaves at least 15V margin
for the leakage spike across line and load conditions. A
larger voltage margin will be required for poorly wound
transformers or for excessive leakage inductance.
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LT8302ES8E#PBF

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Buy LT8302ES8E#PBF
Manufacturer:
Analog Devices / Linear Technology
Description:
Switching Voltage Regulators 42VIN Micropower No-Opto Isolated Flyback Converter with 65V/4.5A Switch
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