LT1505CG-1#PBF Datasheet LT1505CG-1#PBF, LT1505CG#PBF, LT1505CG-1#TRPBF | P10-P12

LT1505

1505fc

The LT1505 is a synchronous current mode PWM step-

down (buck) switcher. The battery DC charge current is pro-

grammed by a resistor R

PROG

(or a DAC output current) at

the PROG pin and the ratio of sense resistors R

over R

(see Block Diagram). Amplifier CA1 converts the charge cur-

rent through R

to a much lower current I

PROG

BAT

• RS1/RS2) fed into the PROG pin. Amplifier CA2 com-

pares the output of CA1 with the programmed current and

drives the PWM loop to force them to be equal. High DC

accuracy is achieved with averaging capacitor C

PROG

. Note

that I

PROG

has both AC and DC components. I

PROG

goes

through R1 and generates a ramp signal that is fed to the

PWM control comparator C1 through buffer B1 and level

shift resistors R2 and R3, forming the current mode inner

loop. The BOOST pin supplies the topside power switch gate

drive. The LT1505 generates an 9.1V V

GBIAS

to power drives

and V

BOOSTC

. BOOSTC pin supplies the current amplifier

CA1 with a voltage higher than V

for low dropout appli-

cation. For batteries like lithium that require both constant-

current and constant-voltage charging, the 0.5% 2.465V

reference and the amplifier VA reduce the charge current

when battery voltage reaches the preset level. For NiMH and

NiCd, VA can be used for overvoltage protection.

The amplifier CL1 monitors and limits the input current,

normally from the AC adapter, to a preset level (92mV/R

At input current limit, CL1 will supply the programming

current I

PROG

, thus reducing battery charging current.

To prevent current shoot-through between topside and

lowside switches, comparators A3 and A4 assure that one

switch turns off before the other is allowed to turn on.

Comparator A12 monitors charge current level and turns

lowside switch off if it drops below 20% of the programmed

value (20mV across R

) to allow for inductor discontinu-

ous mode operation. Therefore sometimes even in con-

tinuous mode operation with light current level the lowside

switch stays off.

Comparator E6 monitors the charge current and signals

through the FLAG pin when the charger is in voltage mode

and the charge current level is reduced to 20%. This charge

complete signal can be used to start a timer for charge

termination.

The INFET pin drives an external P-channel FET for low

dropout application.

When input voltage is removed, V

will be held up by the

body diode of the topside MOSFET. The LT1505 goes into

a low current, 10µA typical, sleep mode as V

drops

below the battery voltage. To shut down the charger

simply pull the V

pin or SHDN pin low with a transistor.

OPERATION

APPLICATIONS INFORMATION

WUU

Input and Output Capacitors

In the 4A Lithium Battery Charger (Figure 1), the input

capacitor (C

) is assumed to absorb all input switching

ripple current in the converter, so it must have adequate

ripple current rating. Worst-case RMS ripple current will

be equal to one half of output charging current. Actual

capacitance value is not critical. Solid tantalum capacitors

such as the AVX TPS and Sprague 593D series have high

ripple current rating in a relatively small surface mount

package, but

caution must be used when tantalum capaci-

tors are used for input bypass

. High input surge currents

can be created when the adapter is hot-plugged to the

charger and solid tantalum capacitors have a known

failure mechanism when subjected to very high turn-on

surge currents. Highest possible voltage rating on the

capacitor will minimize problems. Consult the manufac-

turer before use. Alternatives include new high capacity

ceramic (at least 20µF) from Tokin or United Chemi-Con/

Marcon, et al.

The output capacitor (C

OUT

) is also assumed to absorb

output switching current ripple. The general formula for

capacitor current is:

RMS

(L1)(f)

BAT

()

0.29 (V

BAT

) 1 –

For example, V

= 19V, V

BAT

= 12.6V, L1 = 15µH,

and f = 200kHz, I

RMS

= 0.4A.

LT1505

1505fc

APPLICATIONS INFORMATION

WUU

EMI considerations usually make it desirable to minimize

ripple current in the battery leads. Beads or inductors may

be added to increase battery impedance at the 200kHz

switching frequency. Switching ripple current splits be-

tween the battery and the output capacitor depending on

the ESR of the output capacitor and the battery imped-

ance. If the ESR of C

OUT

is 0.2Ω and the battery impedance

is raised to 4Ω with a bead or inductor, only 5% of the

ripple current will flow in the battery.

Soft Start and Undervoltage Lockout

The LT1505 is soft started by the 0.33µF capacitor on the

pin. On start-up, the V

pin voltage will rise quickly to

0.5V, then ramp up at a rate set by the internal 45µA pull-

up current and the external capacitor. Battery charge

current starts ramping up when V

voltage reaches 0.7V

and full current is achieved with V

at 1.1V. With a 0.33µF

capacitor, time to reach full charge current is about 10ms

and it is assumed that input voltage to the charger will

reach full value in less than 10ms. The capacitor can be

increased up to 1µF if longer input start-up times are

needed.

In any switching regulator, conventional timer-based soft

starting can be defeated if the input voltage rises much

slower than the time out period. This happens because the

switching regulators in the battery charger and the com-

puter power supply are typically supplying a fixed amount

of power to the load. If input voltage comes up slowly

compared to the soft start time, the regulators will try to

deliver full power to the load when the input voltage is still

well below its final value. If the adapter is current limited,

it cannot deliver full power at reduced output voltages and

the possibility exists for a quasi “latch” state where the

adapter output stays in a current limited state at reduced

output voltage. For instance, if maximum charger plus

computer load power is 30W, a 15V adapter might be

current limited at 2.5A. If adapter voltage is less than

(30W/2.5A = 12V) when full power is drawn, the adapter

voltage will be pulled down by the constant 30W load until

it reaches a lower stable state where the switching regu-

lators can no longer supply full load. This situation can be

prevented by setting

undervoltage lockout

higher than the

minimum adapter voltage where full power can be achieved.

Figure 2. Adapter Current Limiting

92mV

–

500Ω

CLP

CLN

1505 F02

LT1505

1µF

CL1

AC ADAPTER

OUTPUT

92mV

ADAPTER CURRENT LIMIT

A resistor divider is used to set the desired V

lockout

voltage as shown in Figure 2. A typical value for R6 is 5k

and R5 is found from:

R5 =

R6(V – V )

= Rising lockout threshold on the UV pin

= Charger input voltage that will sustain full load power

Example: With R6 = 5k, V

= 6.7V and setting V

at 16V;

R5 = 5k (16V – 6.7V)/6.7V = 6.9k

The resistor divider should be connected directly to the

adapter output as shown, not to the V

pin to prevent

battery drain with no adapter voltage. If the UV pin is not

used, connect it to the adapter output (not V

) and

connect a resistor no greater than 5k to ground. Floating

the pin will cause reverse battery current to increase from

10µA to 200µA.

Adapter Current Limiting

(Not Applicable for the LT1505-1)

An important feature of the LT1505 is the ability to

automatically adjust charge current to a level which avoids

overloading the wall adapter. This allows the product to

operate at the same time batteries are being charged

without complex load management algorithms. Addition-

ally, batteries will automatically be charged at the maximum

possible rate of which the adapter is capable.

LT1505

1505fc

APPLICATIONS INFORMATION

WUU

This is accomplished by sensing total adapter output

current and adjusting charge current downward if a preset

adapter current limit is exceeded. True analog control is

used, with closed loop feedback ensuring that adapter load

current remains within limits. Amplifier CL1 in Figure 2

senses the voltage across R

, connected between the

CLP and CLN pins. When this voltage exceeds 92mV, the

amplifier will override programmed charge current to limit

adapter current to 92mV/R

. A lowpass filter formed by

500Ω and 1µF is required to eliminate switching noise. If

the current limit is not used, then the R7 /C1 filter and the

COMP1 (R1/C7) compensation networks are not needed,

and both CLP and CLN pins should be connected to V

Charge Current Programming

The basic formula for charge current is (see Block

Diagram):

BAT

= I

PROG

2.465V

PROG

()()

()

where R

PROG

is the total resistance from PROG pin to ground.

For the sense amplifier CA1 biasing purpose, R

should

have the same value as R

and SPIN should be connected

directly to the sense resistor (R

) as shown in the Block

Diagram.

For example, 4A charging current is needed. For low power

dissipation on R

and enough signal to drive the amplifier

CA1, let R

= 100mV/4A = 0.025Ω. This limits R

power

to 0.4W. Let R

PROG

= 5k, then:

= R

= = 200Ω

BAT

)(R

PROG

)(R

)

2.465V

(4A)(5k)(0.025)

2.465V

Charge current can also be programmed by pulse width

modulating I

PROG

with a switch Q1 to R

PROG

at a frequency

higher than a few kHz (Figure 3). Charge current will be

proportional to the duty cycle of the switch with full current

at 100% duty cycle.

When a microprocessor DAC output is used to control

charge current, it must be capable of sinking current at a

compliance up to 2.5V if connected directly to the PROG

pin.

Note that for charge current accuracy and noise immu-

nity, 100mV full scale level across the sense resistor RS1

is required. Consequently, both RS2 and RS3 should be

200Ω.

It is critical to have a good Kelvin connection on the

current sense resistor RS1 to minimize stray resistive

and inductive pickup. RS1 should have low parasitic

inductance (typical 3nH or less, as exhibited by Dale or

IRC sense resistors). The layout path from RS2 and RS3

to RS1 should be kept away from the fast switching SW

node. Under low charge current conditions, a low quality

sense resistor with high ESL (4nH or higher) coupled

with a very noisy current sense path might false trip

comparator A12 and turn on BGATE at the wrong time,

potentially damaging the bottom power FET. In this case,

an RC filter of 10Ω and 10nF should be used across RS1

to filter out the noise (see Figure 4).

PWM

PROG

4.7k

PROG

1µF

VN2222

LT1505

1505 F03

BAT

= (DC)(4A)

Figure 3. PWM Current Programming

Figure 4. Reducing Current Sensing Noise

1505 F04

LT1505

SPIN

SENSE

BAT

BAT2

–

RS1

RS2

RS3

10Ω

10nF

+ V

RS1

–

BATTERY

Lithium-Ion Charging

The 4A Lithium Battery Charger (Figure 1) charges lithium-

ion batteries at a constant 4A until battery voltage reaches

the preset value. The charger will then automatically go

into a constant-voltage mode with current decreasing to

near zero over time as the battery reaches full charge.

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LT1505CG-1#PBF

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Manufacturer:

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Description:

Battery Management Const-C/V Hi Eff Bat Chr

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LT1505CG-1#PBF

LT1505CG-1#PBF

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