Data Sheet AD9631/AD9632
Rev. D | Page 15 of 20
THEORY OF OPERATION
GENERAL
The AD9631/AD9632 are wide bandwidth, voltage feedback
amplifiers. Because their open-loop frequency response follows
the conventional 6 dB/octave roll-off, their gain bandwidth
product is basically constant. Increasing their closed-loop gain
results in a corresponding decrease in small signal bandwidth.
This can be observed by noting the bandwidth specification
between the AD9631 (gain of +1) and AD9632 (gain of +2). The
AD9631/AD9632 typically maintain 65° of phase margin. This
high margin minimizes the effects of signal and noise peaking.
FEEDBACK RESISTOR CHOICE
The value of the feedback resistor is critical for optimum per-
formance on the AD9631 (gain of +1) and less critical as the
gain increases. Therefore, this section is specifically targeted
at the AD9631.
At the minimum stable gain (+1), the AD9631 provides opti-
mum dynamic performance with R
F
= 140 Ω. This resistor acts
as a parasitic suppressor only against damped RF oscillations
that can occur due to lead (input, feedback) inductance and
parasitic capacitance. This value of R
F
provides the best combi-
nation of wide bandwidth, low parasitic peaking, and fast
settling time.
In fact, for the same reasons, place a 100 Ω to 130 Ω resistor in
series with the positive input for other AD9631 noninverting
and all AD9631 inverting configurations. The correct connec-
tion is shown in Figure 59 and Figure 60.
Figure 59. Noninverting Operation
Figure 60. Inverting Operation
When the AD9631 is used in the transimpedance (I to V)
mode, such as in photodiode detection, the value of R
F
and
diode capacitance (C
I
) are usually known. Generally, the value
of R
F
selected will be in the kΩ range, and a shunt capacitor (C
F
)
across R
F
will be required to maintain good amplifier stability.
The value of C
F
required to maintain optimal flatness (<1 dB
peaking) and settling time can be estimated by
( )
[ ]
2
1
22
/12
F
O
FI
O
F
RRCC
ωω
−≅
where:
ω
O
is equal to the unity gain bandwidth product of the amplifier
in rad/sec.
C
I
is the equivalent total input capacitance at the inverting input.
Typically ω
O
= 800 × 10
6
rad/sec (see Figure 19).
As an example, choosing R
F
= 10 kΩ and C
I
= 5 pF requires C
F
to be 1.1 pF (Note that C
I
includes both source and parasitic
circuit capacitance). The bandwidth of the amplifier can be
estimated using C
F
:
Figure 61. Transimpedance Configuration
For general voltage gain applications, the amplifier bandwidth
can be closely estimated as
( )
G
F
O
3dB
RR
f
/12 +
≅
π
ω
This estimation loses accuracy for gains of +2/−1 or lower due
to the damping factor of the amplifier. For these low gain cases,
the bandwidth will actually extend beyond the calculated value
(see Figure 17 and Figure 29).
As a general rule, Capacitor C
F
will not be required if
( )
O
I
G
F
NG
CRR
ω
4
≤×
where NG is the noise gain (1 + R
F
/R
G
) of the circuit. For most
voltage gain applications, this should be the case.
+V
S
0.1µF
0.1µF
10µF
10µF
–V
S
100Ω TO
130Ω
R
IN
R
TERM
V
IN
V
OUT
R
F
R
G
R
F
R
G
G = 1 +
AD9631/
AD9632
00601-059
+V
S
0.1µF
0.1µF
10µF
10µF
–V
S
R
TERM
V
IN
V
OUT
R
F
R
G
R
F
R
G
G = 1 –
AD9631/
AD9632
100Ω TO
130Ω
R
IN
00601-060
V
OUT
R
F
C
F
C
I
I
I
AD9631
00601-061
