AD600/AD602
Rev. F | Page 18 of 32
FREQUENCY (MHz)
AGC OUTPUT CHANGE (dB)
1
100
10
0.1
A simple half-wave detector is used based on Q1 and R2. The
average current into Capacitor C2 is the difference between the
current provided by the AD590 (300 µA at 300 K, 27°C) and the
collector current of Q1. In turn, the control voltage, V
G
, is the
time integral of this error current. When V
G
(thus the gain) is
stable, the rectified current in Q1 must, on average, balance
exactly the current in the AD590. If the output of A2 is too small
to do this, V
G
ramps up, causing the gain to increase until Q1
conducts sufficiently. The operation of this control system follows.
First, consider the particular case where R2 is zero and the
output voltage, V
OUT
, is a square wave at, for example, 100 kHz,
well above the corner frequency of the control loop. During the
time V
OUT
is negative, Q1 conducts. When V
OUT
is positive, it is
cut off. Because the average collector current is forced to be
300 A and the square wave has a 50% duty-cycle, the current
when conducting must be 600 A. With R2 omitted, the peak
value of V
OUT
would be just the V
BE
of Q1 at 600 A (typically
about 700 mV) or 2 V
BE
p-p. This voltage, thus the amplitude at
which the output stabilizes, has a strong negative temperature
coefficient (TC), typically –1.7 mV/°C. While this may not be
troublesome in some applications, the correct value of R2
renders the output stable with temperature.
To understand this, first note that the current in the AD590 is
closely proportional to absolute temperature (PTAT). In fact,
this IC is intended for use as a thermometer. For the moment,
assume that the signal is a square wave. When Q1 is conducting,
V
OUT
is the sum of V
BE
and a voltage that is PTAT and that can
be chosen to have an equal but opposite TC of the base-to-
emitter voltage. This is actually nothing more than the band gap
voltage reference principle thinly disguised. When R2 is chosen
so that the sum of the voltage across it and the V
BE
of Q1 is close
to the band gap voltage of about 1.2 V, V
OUT
is stable over a wide
range of temperatures, provided that Q1 and the AD590 share the
same thermal environment.
Because the average emitter current is 600 A during each half-
cycle of the square wave, a resistor of 833 would add a PTAT
voltage of 500 mV at 300 K, increasing by 1.66 mV/°C. In
practice, the optimum value of R2 depends on the transistor
used and, to a lesser extent, on the waveform for which the
temperature stability is to be optimized; for the devices shown
and sine wave signals, the recommended value is 806 . This
resistor also serves to lower the peak current in Q1, and the
200 Hz LP filter it forms with C2 helps to minimize distortion
due to ripple in V
G
. Note that the output amplitude under sine
wave conditions is higher than for a square wave because the
average value of the current for an ideal rectifier would be
0.637 times as large, causing the output amplitude to be 1.88 V
(= 1.2/0.637), or 1.33 V rms. In practice, the somewhat nonideal
rectifier results in the sine wave output being regulated to about
1.275 V rms.
An offset of 375 mV is applied to the inverting gain-control
inputs C1LO and C2LO. Therefore, the nominal –625 mV to
+625 mV range for V
G
is translated upward (at V
G
´) to –0.25 V
for minimum gain to +1 V for maximum gain. This prevents
Q1 from going into heavy saturation at low gains and leaves
sufficient headroom of 4 V for the AD590 to operate correctly
at high gains when using a 5 V supply.
In fact, the 6 dB interstage attenuator means that the overall
gain of this AGC system actually runs from –6 dB to +74 dB.
Thus, an input of 2 V rms would be required to produce a
1 V rms output at the minimum gain, which exceeds the 1 V rms
maximum input specification of the AD600. The available gain
range is therefore 0 dB to 74 dB (or X1 to X5000). Because the
gain scaling is 15.625 mV/dB (because of the cascaded stages),
the minimum value of V
G
´ is actually increased by 6 × +15.625 mV,
or about 94 mV, to −156 mV, so the risk of saturation in Q1 is
reduced.
The emitter circuit of Q1 is somewhat inductive (due to its
finite f
t
and base resistance). Consequently, the effective value of
R2 increases with frequency. This results in an increase in the
stabilized output amplitude at high frequencies, but for the
addition of C3, determined experimentally to be 15 pF for the
2N3904 for maximum response flatness. Alternatively, a faster
transistor can be used here to reduce HF peaking. Figure 38
shows the ac response at the stabilized output level of about
1.3 rms. Figure 39 demonstrates the output stabilization for the
sine wave inputs of 1 mV rms to 1 V rms at frequencies of 100 kHz,
1 MHz, and 10 MHz.
3dB
00538-036
Figure 38. AC Response at the Stabilized Output Level of 1.3 V rms
