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LTC3783IFE#TRPBF

P1-P3P4-P6P7-P9P10-P12P13-P15P16-P18P19-P21P22-P24
LTC3783
16
3783fb
OPERATION
During the switch on-time, the IMAX comparator limits the
absolute maximum voltage drop across the power MOSFET
to a nominal 150mV, regardless of duty cycle. The peak
inductor current is therefore limited to 150mV/R
DS(ON)
.
The relationship between the maximum load current, duty
cycle, and the R
DS(ON)
of the power MOSFET is:
R
DS(ON)
< 150mV •
1–D
MAX
1+
c
2
•I
OUT(MAX)
• r
T
The r
T
term accounts for the temperature coefficient of
the R
DS(ON)
of the MOSFET, which is typically 0.4%/°C.
Figure 8 illustrates the variation of normalized R
DS(ON)
over temperature for a typical power MOSFET.
It is worth noting that the 1 - D
MAX
relationship between
I
O(MAX)
and R
DS(ON)
can cause boost converters with a
wide input range to experience a dramatic range of maxi-
mum input and output currents. This should be taken into
consideration in applications where it is important to limit
the maximum current drawn from the input supply, and
also to avoid triggering the 150mV IMAX comparator, as
this condition can result in excessive noise.
Calculating Power MOSFET Switching and Conduction
Losses and Junction Temperatures
In order to calculate the junction temperature of the power
MOSFET, the power dissipated by the device must be known.
This power dissipation is a function of the duty cycle, the
load current, and the junction temperature itself (due to
the positive temperature coefficient of its R
DS(ON)
. As a
result, some iterative calculation is normally required to
determine a reasonably accurate value. Since the controller
is using the MOSFET as both a switching and a sensing
element, care should be taken to ensure that the converter
is capable of delivering the required load current over all
operating conditions (line voltage and temperature), and
for the worst-case specifications for V
SENSE(MAX)
and the
R
DS(ON
) of the MOSFET listed in the manufacturer’s data
sheet.
The power dissipated by the MOSFET in a boost converter
is:
P
FET
=
I
OUT(MAX)
1–D
MAX
2
•R
DS(ON)
•D
MAX
• r
T
+
k • V
OUT
1.85
•
I
OUT(MAX)
1–D
MAX
• C
RSS
• f
The first term in the equation above represents the I
2
R
losses in the device, and the second term, the switching
losses. The constant k = 1.7 is an empirical factor inversely
related to the gate drive current and has the dimension
of 1/current.
JUNCTION TEMPERATURE (°C)
–50
ρ
T
NORMALIZED ON RESISTANCE
1.0
1.5
150
3783 F08
0.5
0
0
50
100
2.0
Figure 8. Normalized R
DS(ON)
vs Temperature
Another method of choosing which power MOSFET to
use is to check what the maximum output current is for a
given R
DS(ON)
, since MOSFET on-resistances are available
in discrete values.
I
O(MAX)
= 150mV •
1– D
MAX
1+
c
2
• R
DS(ON)
• r
T
LTC3783
17
3783fb
OPERATION
From a known power dissipated in the power MOSFET, its
junction temperature can be obtained using the following
formula:
T
J
= T
A
+ P
FET
• θ
JA
The θ
JA
to be used in this equation normally includes the
θ
JC
for the device plus the thermal resistance from the
case to the ambient temperature (
θ
CA
). This value of T
J
can then be compared to the original, assumed value used
in the iterative calculation process.
Boost Converter: Output Diode Selection
To maximize efficiency, a fast switching diode with low
forward drop and low reverse leakage is desired. The output
diode in a boost converter conducts current during the
switch off-time. The peak reverse voltage that the diode
must withstand is equal to the regulator output voltage.
The average forward current in normal operation is equal
to the output current, and the peak current is equal to the
peak inductor current.
I
D(PEAK)
= I
L(PEAK)
= 1+
c
2
•
I
OUT(MAX)
1– D
MAX
The power dissipated by the diode is:
P
D
= I
OUT(MAX)
• V
D
and the diode junction temperature is:
T
J
= T
A
+ P
D
• θ
JA
The θ
JA
to be used in this equation normally includes the
θ
JC
for the device plus the thermal resistance from the
board to the ambient temperature in the enclosure.
Remember to keep the diode lead lengths short and to
observe proper switch-node layout (see Board Layout
Checklist) to avoid excessive ringing and increased
dissipation.
Boost Converter: Output Capacitor Selection
Contributions of ESR (equivalent series resistance), ESL
(equivalent series inductance) and the bulk capacitance
must be considered when choosing the correct component
for a given output ripple voltage. The effects of these three
parameters (ESR, ESL and bulk C) on the output voltage
ripple waveform are illustrated in Figure 9 for a typical
boost converter.
V
OUT
(AC)
∆V
ESR
RINGING DUE TO
TOTAL INDUCTANCE
(BOARD + CAP)
∆V
COUT
3783 F09
Figure 9. Output Ripple Voltage
The choice of component(s) begins with the maximum
acceptable ripple voltage (expressed as a percentage of
the output voltage), and how this ripple should be divided
between the ESR step and the charging/discharging ∆V.
For the purpose of simplicity we will choose 2% for the
maximum output ripple, to be divided equally between the
ESR step and the charging/discharging ∆V. This percentage
ripple will change, depending on the requirements of the
application, and the equations provided below can easily
be modified.
For a 1% contribution to the total ripple voltage, the ESR
of the output capacitor can be determined using the fol-
lowing equation:
ESR
COUT
< 0.01•
V
OUT
I
IN(PEAK)
where :
I
IN(PEAK)
= 1+
c
2
•
I
OUT(MAX)
1– D
MAX
For the bulk C component, which also contributes 1% to
the total ripple:
C
OUT
>
I
OUT(MAX)
0.01• V
OUT
• f
LTC3783
18
3783fb
OPERATION
For many designs it is possible to choose a single capacitor
type that satisfies both the ESR and bulk C requirements
for the design. In certain demanding applications, however,
the ripple voltage can be improved significantly by con-
necting two or more types of capacitors in parallel. For
example, using a low ESR ceramic capacitor can minimize
the ESR setup, while an electrolytic capacitor can be used
to supply the required bulk C.
Once the output capacitor ESR and bulk capacitance have
been determined, the overall ripple voltage waveform
should be verified on a dedicated PC board (see Board
Layout section for more information on component place-
ment). Lab breadboards generally suffer from excessive
series inductance (due to inter-component wiring), and
these parasitics can make the switching waveforms look
significantly worse than they would be on a properly
designed PC board.
The output capacitor in a boost regulator experiences
high RMS ripple currents. The RMS output capacitor
ripple current is:
I
RMS(COUT)
; I
OUT(MAX)
•
V
OUT
– V
IN(MIN)
V
IN(MIN)
Note that the ripple current ratings from capacitor manu-
facturers are often based on only 2000 hours of life. This
makes it advisable to further derate the capacitor or to
choose a capacitor rated at a higher temperature than
required. Several capacitors may also be placed in parallel
to meet size or height requirements in the design.
Boost Converter: Input Capacitor Selection
The input capacitor of a boost converter is less critical
than the output capacitor, due to the fact that the inductor
is in series with the input, and hence, the input current
waveform is continuous (see Figure 10). The input volt-
age source impedance determines the size of the input
capacitor, which is typically in the range of 10µF to 100µF.
A low ESR capacitor is recommended, although it is not
as critical as for the output capacitor.
I
IN
I
L
3783 F10
Figure 10. Inductor and Input Currents
The RMS input capacitor ripple current for a boost
converter is:
I
RMS(CIN)
; 0.3•
V
IN(MIN)
L • f
•D
MAX
Please note that the input capacitor can see a very high
surge current when a battery is suddenly connected to
the input of the converter, and solid tantalum capacitors
can fail catastrophically under these conditions. Be sure
to specify surge-tested capacitors!
Boost Converter Design Example
The design example given here will be for the circuit shown
in Figure 1. The input voltage is 12V, and the output voltage
is 25V at a maximum load current of 0.7A (1A peak).
1. The duty cycle is:
D=
V
OUT
+ V
D
– V
IN
V
OUT
+ V
D
=
25+ 0.4–12
25+ 0.4
= 53%
2. The operating frequency is chosen to be 1MHz to
maximize the PWM dimming range. From Figure 2, the
resistor from the FREQ pin to ground is 6k.
3. An inductor ripple current of 40% of the maximum load
current is chosen, so the peak input current (which is also
the minimum saturation current) is:
I
IN(PEAK)
= 1+
c
2
•
I
OUT(MAX)
1– D
MAX
= 1.2 •
0.7
1– 0.53
= 1.8A
The inductor ripple current is:
∆I
L
= c •
I
OUT(MAX)
1− D
MAX
= 0.4 •
0.7
1− 0.53
= 0.6A
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LTC3783IFE#TRPBF

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Buy LTC3783IFE#TRPBF
Manufacturer:
Analog Devices / Linear Technology
Description:
Switching Voltage Regulators PWM LED Drvr & Boost, Fly & SEPIC Conv
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