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LTC1778IGN#PBF

P1-P3P4-P6P7-P9P10-P12P13-P15P16-P18P19-P21P22-P24
10
LTC1778/LTC1778-1
1778fb
APPLICATIO S I FOR ATIO
WUUU
The basic LTC1778 application circuit is shown in
Figure 1. External component selection is primarily de-
termined by the maximum load current and begins with
the selection of the sense resistance and power MOSFET
switches. The LTC1778 uses the on-resistance of the
synchronous power MOSFET for determining the induc-
tor current. The desired amount of ripple current and
operating frequency largely determines the inductor value.
Finally, C
IN
is selected for its ability to handle the large
RMS current into the converter and C
OUT
is chosen with
low enough ESR to meet the output voltage ripple and
transient specification.
Choosing the LTC1778 or LTC1778-1
The LTC1778 has an open-drain PGOOD output that
indicates when the output voltage is within ±7.5
% of the
regulation point. The LTC1778-1 trades the PGOOD pin for
a V
ON
pin that allows the on-time to be adjusted. Tying the
V
ON
pin high results in lower values for R
ON
which is useful
in high V
OUT
applications. The V
ON
pin also provides a
means to adjust the on-time to maintain constant fre-
quency operation in applications where V
OUT
changes and
to correct minor frequency shifts with changes in load
current. Finally, the V
ON
pin can be used to provide
additional current limiting in positive-to-negative convert-
ers and as a control input to synchronize the switching
frequency with a phase locked loop.
Maximum Sense Voltage and V
RNG
Pin
Inductor current is determined by measuring the voltage
across a sense resistance that appears between the PGND
and SW pins. The maximum sense voltage is set by the
voltage applied to the V
RNG
pin and is equal to approxi-
mately (0.133)V
RNG
. The current mode control loop will
not allow the inductor current valleys to exceed
(0.133)V
RNG
/R
SENSE
. In practice, one should allow some
margin for variations in the LTC1778 and external compo-
nent values and a good guide for selecting the sense
resistance is:
R
V
I
SENSE
RNG
OUT MAX
=
10 •
()
An external resistive divider from INTV
CC
can be used to
set the voltage of the V
RNG
pin between 0.5V and 2V
resulting in nominal sense voltages of 50mV to 200mV.
Additionally, the V
RNG
pin can be tied to SGND or INTV
CC
in which case the nominal sense voltage defaults to 70mV
or 140mV, respectively. The maximum allowed sense
voltage is about 1.33 times this nominal value.
Power MOSFET Selection
The LTC1778 requires two external N-channel power
MOSFETs, one for the top (main) switch and one for the
bottom (synchronous) switch. Important parameters for
the power MOSFETs are the breakdown voltage V
(BR)DSS
,
threshold voltage V
(GS)TH
, on-resistance R
DS(ON)
, reverse
transfer capacitance C
RSS
and maximum current I
DS(MAX)
.
The gate drive voltage is set by the 5V INTV
CC
supply.
Consequently, logic-level threshold MOSFETs must be
used in LTC1778 applications. If the input voltage is
expected to drop below 5V, then sub-logic level threshold
MOSFETs should be considered.
When the bottom MOSFET is used as the current sense
element, particular attention must be paid to its on-
resistance. MOSFET on-resistance is typically specified
with a maximum value R
DS(ON)(MAX)
at 25°C. In this case,
additional margin is required to accommodate the rise in
MOSFET on-resistance with temperature:
R
R
DS ON MAX
SENSE
T
()( )
=
ρ
The ρ
T
term is a normalization factor (unity at 25°C)
accounting for the significant variation in on-resistance
Figure 2. R
DS(ON)
vs. Temperature
JUNCTION TEMPERATURE (°C)
–50
ρ
T
NORMALIZED ON-RESISTANCE
1.0
1.5
150
1778 F02
0.5
0
0
50
100
2.0
11
LTC1778/LTC1778-1
1778fb
APPLICATIO S I FOR ATIO
WUUU
with temperature, typically about 0.4%/°C as shown in
Figure 2. For a maximum junction temperature of 100°C,
using a value ρ
T
= 1.3 is reasonable.
The power dissipated by the top and bottom MOSFETs
strongly depends upon their respective duty cycles and
the load current. When the LTC1778 is operating in
continuous mode, the duty cycles for the MOSFETs are:
D
V
V
D
VV
V
TOP
OUT
IN
BOT
IN OUT
IN
=
=
–
The resulting power dissipation in the MOSFETs at maxi-
mum output current are:
P
TOP
= D
TOP
I
OUT(MAX)
2
ρ
T(TOP)
R
DS(ON)(MAX)
+ k V
IN
2
I
OUT(MAX)
C
RSS
f
P
BOT
= D
BOT
I
OUT(MAX)
2
ρ
T(BOT)
R
DS(ON)(MAX)
Both MOSFETs have I
2
R losses and the top MOSFET
includes an additional term for transition losses, which are
largest at high input voltages. The constant k = 1.7A
–1
can
be used to estimate the amount of transition loss. The
bottom MOSFET losses are greatest when the bottom duty
cycle is near 100%, during a short-circuit or at high input
voltage.
Operating Frequency
The choice of operating frequency is a tradeoff between
efficiency and component size. Low frequency operation
improves efficiency by reducing MOSFET switching losses
but requires larger inductance and/or capacitance in order
to maintain low output ripple voltage.
The operating frequency of LTC1778 applications is deter-
mined implicitly by the one-shot timer that controls the
on-time t
ON
of the top MOSFET switch. The on-time is set
by the current into the I
ON
pin and the voltage at the V
ON
pin (LTC1778-1) according to:
t
V
I
pF
ON
VON
ION
= ()10
V
ON
defaults to 0.7V in the LTC1778.
Tying a resistor R
ON
from V
IN
to the I
ON
pin yields an on-
time inversely proportional to V
IN
. For a step-down con-
verter, this results in approximately constant frequency
operation as the input supply varies:
f
V
VR pF
H
OUT
VON ON
Z
=
()
[]
10
To hold frequency constant during output voltage changes,
tie the V
ON
pin to V
OUT
or to a resistive divider from V
OUT
when V
OUT
> 2.4V. The V
ON
pin has internal clamps that
limit its input to the one-shot timer. If the pin is tied below
0.7V, the input to the one-shot is clamped at 0.7V. Simi-
larly, if the pin is tied above 2.4V, the input is clamped at
2.4V. In high V
OUT
applications, tying V
ON
to INTV
CC
so
that the comparator input is 2.4V results in a lower value
for R
ON
. Figures 3a and 3b show how R
ON
relates to
switching frequency for several common output voltages.
R
ON
(kΩ)
100
100
SWITCHING FREQUENCY (kHz)
1000
1000 10000
1778 F03a
V
OUT
= 3.3V
V
OUT
= 1.5V
V
OUT
= 2.5V
R
ON
(kΩ)
100
100
SWITCHING FREQUENCY (kHz)
1000
1000 10000
1778 F03b
V
OUT
= 3.3V
V
OUT
= 12V
V
OUT
= 5V
Figure 3a. Switching Frequency vs R
ON
for the LTC1778 and LTC1778-1 (V
ON
= 0V)
Figure 3b. Switching Frequency vs R
ON
for the LTC1778-1 (V
ON
= INTV
CC
)
12
LTC1778/LTC1778-1
1778fb
Because the voltage at the I
ON
pin is about 0.7V, the
current into this pin is not exactly inversely proportional to
V
IN
, especially in applications with lower input voltages.
To correct for this error, an additional resistor R
ON2
connected from the I
ON
pin to the 5V INTV
CC
supply will
further stabilize the frequency.
R
V
V
R
ON ON2
5
07
=
.
Changes in the load current magnitude will also cause
frequency shift. Parasitic resistance in the MOSFET
switches and inductor reduce the effective voltage across
the inductance, resulting in increased duty cycle as the
load current increases. By lengthening the on-time slightly
as current increases, constant frequency operation can be
maintained. This is accomplished with a resistive divider
from the I
TH
pin to the V
ON
pin and V
OUT
. The values
required will depend on the parasitic resistances in the
specific application. A good starting point is to feed about
25% of the voltage change at the I
TH
pin to the V
ON
pin as
shown in Figure 4a. Place capacitance on the V
ON
pin to
filter out the I
TH
variations at the switching frequency. The
resistor load on I
TH
reduces the DC gain of the error amp
and degrades load regulation, which can be avoided by
using the PNP emitter follower of Figure 4b.
Minimum Off-time and Dropout Operation
The minimum off-time t
OFF(MIN)
is the smallest amount of
time that the LTC1778 is capable of turning on the bottom
MOSFET, tripping the current comparator and turning the
MOSFET back off. This time is generally about 250ns. The
minimum off-time limit imposes a maximum duty cycle of
t
ON
/(t
ON
+ t
OFF(MIN)
). If the maximum duty cycle is reached,
due to a dropping input voltage for example, then the
output will drop out of regulation. The minimum input
voltage to avoid dropout is:
VV
tt
t
IN MIN OUT
ON OFF MIN
ON
()
()
=
+
A plot of maximum duty cycle vs frequency is shown in
Figure 5.
Inductor Selection
Given the desired input and output voltages, the inductor
value and operating frequency determine the ripple
current:
∆ =
⎛
⎝
⎜
⎞
⎠
⎟
−
⎛
⎝
⎜
⎞
⎠
⎟
I
V
fL
V
V
L
OUT OUT
IN
1
Lower ripple current reduces core losses in the inductor,
ESR losses in the output capacitors and output voltage
APPLICATIO S I FOR ATIO
WUUU
C
VON
0.01µF
R
VON2
100k
R
VON1
30k
C
C
V
OUT
R
C
(4a) (4b)
V
ON
I
TH
LTC1778
C
VON
0.01µF
R
VON2
10k
Q1
2N5087
R
VON1
3k
10k
C
C
1778 F04
V
OUT
INTV
CC
R
C
V
ON
I
TH
LTC1778
Figure 4. Correcting Frequency Shift with Load Current Changes
2.0
1.5
1.0
0.5
0
0 0.25 0.50 0.75
1778 F05
1.0
DROPOUT
REGION
DUTY CYCLE (V
OUT
/V
IN
)
SWITCHING FREQUENCY (MHz)
Figure 5. Maximum Switching Frequency vs Duty Cycle
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LTC1778IGN#PBF

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Manufacturer:
Analog Devices / Linear Technology
Description:
Switching Voltage Regulators Synchronous Step down Controller
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