IRS2982STRPBF Datasheet IRS2982STRPBF | P13-P15

IRS2982SPBF

13 2017-02-07

Internal high voltage regulator

The internal high voltage regulator supplies the IC

low voltage bias supply VCC during startup,

allowing operation directly from a DC input voltage

up to 600V. To begin operating the IRS2982

requires VCC to be raised above the under

voltage lockout positive threshold (V

CCUV+

)

and to

continue operating VCC must be maintained

above the under voltage lockout negative

threshold (V

CCUV-

)

The HV regulator enables an IRS2982 based LED

driver to start up very rapidly and deliver light

within 0.5s of switch on at any line input voltage.

When the switching converter is operating VCC is

normally supplied through an auxiliary transformer

winding. The HV regulator switches over to

support mode when steady state operation is

reached in which VCC is held above V

CCUV+

maintain operation under light load or fault

conditions.

As well as supplying VCC the Flyback

inductor/transformer auxiliary winding provides

output voltage and zero-crossing (ZX) information

for critical conduction mode (CrCM) operation.

In the event of a short circuit at the output, the

VCC supply derived from the auxiliary winding

normally collapses below V

CCUV-

causing the

IRS2982 to shut off. The startup sequence then

begins again in a continuous “hiccup” mode until

the short circuit is removed thereby preventing

damage to the circuit.

Figure 1: HV regulator characteristics

Figure 1 illustrates the characteristics of the high

voltage regulator. At switch on it operates in

startup mode during which current is supplied to

VCC from the HV input connected to the rectified

high voltage bus. The current supplied depends on

the voltage at VCC and gradually falls as VCC

rises until it cuts off completely at V

HVS_OFF1

. During

normal operation when the voltage at the FB input

exceeds VREF for the first time the HV regulator

switches over to support mode, where current is

suppled to VCC only when the voltage drops

below V

HVS_OFF2

, which is close to V

CCUV+

. This

helps to sustain the VCC supply at light loads such

as during dimming. Once in support mode the

IRS2982 will not revert to start-up mode until VCC

drops below V

CCUV-.

Sustained operation of the HV regulator may is

likely to cause heating and should be avoided.

Further information is given in the performance

graphs section.

Voltage/current regulation

The IRS2982 may be operated using either a

voltage or current feedback loop. Examples of

each are shown above in the typical application

diagrams. The feedback voltage is fed to the FB

input of the IC, which is connected to the internal

transconductance error amplifier inverting input.

The non-inverting input is connected to an internal

temperature compensated band-gap voltage

reference (V

REF

) and the output is connected to the

compensation (COMP) output.

The FB input can be derived from a shunt resistor

returning LED load current to the 0V return in a

non-isolated Flyback LED driver to regulate output

current. Alternatively it can be fed by a divider

from the transformer auxiliary winding to provide

voltage regulation in an isolated power supply or a

divider directly from the output in a non-isolated

power supply. Sensing from the auxiliary winding

may require some additional filtering components

and does not provide highly accurate regulation of

the output voltage.

The compensation (COMP) voltage determines

the switching cycle on time for voltage mode

control. Loop compensation is performed by

means of the transconductance error amplifier

using an external capacitor (CCOMP) connected

to 0V to realize an integrator to provide a stable

error voltage used to control the converter on time.

CCOMP is typically 1μF in high power factor single

stage converters.

At light loads if V

COMP

drops below V

COMPOFF

the

IRS2982 operates in burst.

Burst mode operation

Under light load conditions the COMP capacitor is

discharged by the error amplifier reducing V

COMP

Minimum on time is reached just before V

COMP

falls

below V

COMPOFF

. If the output needs to be reduced

0 2 4 6 8 101214161820

HV startup current (mA)

VCC (V)

Startup

mode

Support

mode

IRS2982SPBF

14 2017-02-07

further then V

COMP

is driven below V

COMPOFF

and

the gate drive is disabled. However the HV start-

up cell does not switch from support to start-up

mode in this case.

Gate drive does not start up again until V

COMP

has

risen above this value by V

COMPOFF_HYS

at which

time it will start to switch at minimum on time.

During burst mode operation the on time typically

remains at minimum t

ONMIN

and the off time is

limited to the minimum off time t

OFFMIN

. Switching

frequency under these conditions is around

320kHz, which is the maximum possible for the

IRS2982. The length of each burst and the period

between bursts are determined by the value of

CCOMP in conjunction with the converter output

capacitor value.

The VCC supply is normally maintained through

the auxiliary winding, however if necessary the

high voltage regulator will supply current to VCC to

keep it just above V

CCUV-

tburst

V(t)

VOUT(t)

VCOMP(t)

Figure 2: Burst mode waveforms

Primary current limiting

Primary MOSFET current is sensed through a

shunt resistor (RCS) connected from the source of

the Flyback MOSFET switch to the DC bus return.

This current waveform is a high frequency ramp

rising from zero at the beginning of each switching

cycle to reach a peak level at the point the

MOSFET is switched off and remaining at zero

during the off time.

At very low input voltages the voltage or current

regulation loop would demand a very high peak

current, which may exceed the maximum rating of

the transformer. To prevent saturation from

occurring, the IRS2982 provides cycle by cycle

primary current limiting with a threshold V

CSTH

the CS pin input.

Under low line or fault conditions where the

MOSFET current is abnormally high the gate drive

is switched off after the blanking time t

BLANK

Leading edge blanking is necessary to avoid false

triggering due to the fast high current switch on

transient that occurs at switch on of the MOSFET

resulting from discharge of parasitic capacitances.

)

VOUT

)

VCS(

VCSTH

Figure 3: Cycle by cycle current limiting

The IRS2982 normally operates in critical

conduction mode (CrCM), also known as transition

or boundary mode. The transformer auxiliary

winding provides a signal to the IRS2982 ZX input

that indicates when all of the energy stored in the

inductor has been transferred to the output. This

triggers the start of the next switching cycle. The

auxiliary winding voltage is divided through RZX1

and RZX2 to provide the ZX pin input signal. The

pulse amplitude ZX is approximately proportional

to the secondary output voltage and therefore the

DC output voltage:

)21(

RZXRZXN

VOUTRZXN

VZX

+⋅

⋅⋅

≈

[1]

Where,

= Number of turns on the auxiliary winding

= Number of turns on the secondary winding

VOUT = DC Output Voltage or LED voltage

In reality V

contains high frequency ringing

resulting from leakage inductance and other circuit

parasitics. This ringing may need to be filtered to

provide acceptable output voltage tracking.

When the converter MOSFET switches off, V

transitions positively. The values of RZX1 and

RZX2 must be selected so that this voltage always

IRS2982SPBF

15 2017-02-07

exceeds the V

ZX+

threshold to ensure CrCM

operation. If the IRS2982 is used in a converter

required to drive loads over a range of voltage

such as a constant current regulated LED driver,

needs to exceed V

ZX+

at the minimum load

voltage. If V

does not exceed V

ZX+

the IRS2982

will not operate in CrCM and remains pulsing with

the t

timeout periodically triggering the next

cycle. In normal CrCM when V

exceeds V

ZX+

the

IRS2982 waits until the V

then drops below V

ZX-

again to initiate the next switching cycle. A

capacitor may be added in parallel with RZX2 to

add a small delay. This may be needed to

minimize switching loss by delaying switch on until

the minimum point or “valley” of the drain voltage.

The IRS2982 includes a minimum off time function

so that if the ZX pin input transitions high and low

in less than t

OFFMIN

, the gate drive output will not

transition high again until the end of this period.

This prevents false tripping by ringing at the ZX

input and also limits the converter maximum

switching frequency by entering DCM under

conditions where the off time would otherwise be

unacceptably short. The minimum off time extends

the operating range of the converter allowing

operation down to very low duty cycles. This

enables dimming designs to be implemented as

well as limiting operating frequency to limit

switching losses and prevent overheating of the

circuit magnetics, the MOSFET and snubber

components.

V(t)

VOUT(t)

VZX(t)

Figure 4: Zero crossing detection

The IRS2982 may also be used in DCM by driving the

ZX input from a delay circuit triggered by the falling

edge of the gate drive output. A simple circuit is

added as shown in figure 5. As the gate drive goes

low a pulse of determined length can be generated to

drive ZX forcing the system to operate with a fixed off

time greater than the preset minimum off time.

CIN

MPFC

RPFC

RCS

IC1

CVCC

RZX

QZX

DZXCCOMP

COM

OUT

VCC

COMP

IRS2982

Figure 5: DCM Operation

Over voltage protection

The ZX input is a multi-function input also used for

output over voltage limiting. In a Flyback converter if

the load becomes disconnected the output voltage

can become very high causing rapid damage to

components as well as presenting a possible

electrical hazard. In order to protect against this the

IRS2982 senses the output voltage indirectly

through the ZX input, since its peak voltage tracks

the output voltage. If the ZX input voltage exceeds

OVTH

the gate drive switches off for a t

period

before starting the next cycle. At the same time the

COMP capacitor is discharged below V

COMPOFF

after a delay the next cycle will begin at minimum

duty cycle and ramp up slowly. This protection

scheme allows the load to be “hot” connected and

disconnected from the converter output without risk

of damaging the circuit by high voltages appearing

at the output. Care should be taken however to

avoid damage to LED loads due to output capacitor

discharge. An NTC thermistor at the output may be

needed to limit the current surge. The overvoltage

threshold is set by choosing the values of RZX1 and

RZX2 appropriately, according to the formula:

)21(

RZXN

RZXRZXN

VOVTH

VOUT

⋅

⋅⋅

≈

[2]

The recommended over voltage threshold is 25%

above the normal operating voltage for LED loads.

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IRS2982STRPBF

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