10
FN9177.3
June 25, 2009
Programming the Output Voltage
When the converter is in regulation there will be 600mV from
the FB pin to the GND pin. Connect a two-resistor voltage
divider across the VO pin and the GND pin with the output
node connected to the FB pin. Scale the voltage-divider
network such that the FB pin is 600mV with respect to the
GND pin when the converter is regulating at the desired
output voltage. The output voltage can be programmed from
600mV to 3.3V.
Programming the output voltage is written as:
Where:
-V
OUT
is the desired output voltage of the converter
-V
REF
is the voltage that the converter regulates to
between the FB pin and the GND pin
-R
TOP
is the voltage-programming resistor that connects
from the FB pin to the VO pin. In addition to setting the
output voltage, this resistor is part of the loop
compensation network
-R
BOTTOM
is the voltage-programming resistor that
connects from the FB pin to the GND pin
Beginning with R
TOP
between 1k to 5kcalculating
R
BOTTOM
is written as:
Programming the PWM Switching Frequency
The ISL6269 does not use a clock signal to produce PWM.
The PWM switching frequency f
SW
is programmed by the
resistor R
FSET
that is connected from the FSET pin to the
GND pin. The approximate PWM switching frequency is
written as:
Estimating the value of R
FSET
is written as:
Where:
-f
SW
is the PWM switching frequency
-R
FSET
is the f
SW
programming resistor
- K = 75 x 10
-12
It is recommended that whenever the control loop
compensation network is modified, f
SW
should be checked
for the correct frequency and if necessary, adjust R
FSET
.
Compensation Design
The LC output filter has a double pole at its resonant frequency
that causes the phase to abruptly roll downward. The R
3
modulator used in the ISL6269 makes the LC output filter
resemble a first order system in which the closed loop stability
can be achieved with a Type II compensation network.
Your local Intersil representative can provide a PC-based
tool that can be used to calculate compensation network
component values and help simulate the loop frequency
response. The compensation network consists of the internal
error amplifier of the ISL6269 and the external components R1,
R2, C1, and C2 as well as the frequency setting components
R
FSET
, and C
FSET,
are identified in the schematic Figure 6.
General Application Design Guide
This design guide is intended to provide a high-level
explanation of the steps necessary to create a single-phase
power converter. It is assumed that the reader is familiar with
many of the basic skills and techniques referenced below. In
V
REF
V
OUT
R
BOTTOM
R
TOP
R
BOTTOM
+
---------------------------------------------------
=
(EQ. 5)
R
BOTTOM
V
REF
R
TOP
V
OUT
V
REF
-------------------------------------
=
(EQ. 6)
f
SW
1
KR
FSET
---------------------------
=
(EQ. 7)
R
FSET
1
Kf
SW
------------------
=
(EQ. 8)
GND
PHASE
UG
LG
C
OUT
L
OUT
ISL6269
Q
HIGH_SIDE
R
FSET
C
FSET
Q
LOW_SIDE
EA
+
VIN
VIN
VO
FB
R2
R1
C1
C2
-
FSET
COMP
REF
C
ESR
GATE DRIVERS
DCR
R
3
MODULATOR
VOUT
FIGURE 6. COMPENSATION REFERENCE CIRCUIT
ISL6269
11
FN9177.3
June 25, 2009
addition to this guide, Intersil provides complete reference
designs that include schematics, bills of materials, and
example board layouts.
Selecting the LC Output Filter
The duty cycle of an ideal buck converter is a function of the
input and the output voltage. This relationship is written as:
The output inductor peak-to-peak ripple current is written as:
A typical step-down DC/DC converter will have an I
PP
of
20% to 40% of the maximum DC output load current. The
value of I
PP
is selected based upon several criteria such as
MOSFET switching loss, inductor core loss, and the resistive
loss of the inductor winding. The DC copper loss of the
inductor can be estimated by:
Where I
LOAD
is the converter output DC current.
The copper loss can be significant so attention has to be
given to the DCR selection. Another factor to consider when
choosing the inductor is its saturation characteristics at
elevated temperature. A saturated inductor could cause
destruction of circuit components, as well as nuisance OCP
faults.
A DC/DC buck regulator must have output capacitance
C
OUT
into which ripple current I
PP
can flow. Current I
PP
develops a corresponding ripple voltage V
PP
across C
OUT,
which is the sum of the voltage drop across the capacitor
ESR and of the voltage change stemming from charge
moved in and out of the capacitor. These two voltages are
written as:
and
If the output of the converter has to support a load with high
pulsating current, several capacitors will need to be paralleled
to reduce the total ESR until the required V
PP
is achieved.
The inductance of the capacitor can cause a brief voltage dip
if the load transient has an extremely high slew rate. Low
inductance capacitors constructed with reverse package
geometry are available. A capacitor dissipates heat as a
function of RMS current and frequency. Be sure that I
PP
is
shared by a sufficient quantity of paralleled capacitors so that
they operate below the maximum rated RMS current at f
SW
.
Take into account that the rated value of a capacitor can fade
as much as 50% as the DC voltage across it increases.
Selection of the Input Capacitor
The important parameters for the bulk input capacitance are
the voltage rating and the RMS current rating. For reliable
operation, select bulk capacitors with voltage and current
ratings above the maximum input voltage and capable of
supplying the RMS current required by the switching circuit.
Their voltage rating should be at least 1.25 times greater
than the maximum input voltage, while a voltage rating of 1.5
times is a preferred rating. Figure 7 is a graph of the input
RMS ripple current, normalized relative to output load current,
as a function of duty cycle that is adjusted for converter
efficiency. The ripple current calculation is written as:
Where:
-I
MAX
is the maximum continuous I
LOAD
of the converter
- x is a multiplier (0 to 1) corresponding to the inductor
peak-to-peak ripple amplitude expressed as a
percentage of I
MAX
(0% to 100%)
- D is the duty cycle that is adjusted to take into account
the efficiency of the converter which is written as:
In addition to the bulk capacitance, some low ESL ceramic
capacitance is recommended to decouple between the drain
of the high-side MOSFET and the source of the low-side
MOSFET.
D
V
OUT
V
IN
----------------
=
(EQ. 9)
(EQ. 10)
I
PP
V
OUT
1D
f
SW
L
OUT
--------------------------------------
=
(EQ. 11)
P
COPPER
I
LOAD
2
DCR=
V
ESR
I
PP
E SR=
(EQ. 12)
V
C
I
PP
8C
OUT
f
SW
-------------------------------------
=
(EQ. 13)
(EQ. 14)
I
IN_RMS
I
MAX
2
DD
2
xI
MAX
2
D
12
------



+
I
MAX
-----------------------------------------------------------------------------------------------------
=
D
V
OUT
V
IN
EFF
--------------------------
=
(EQ. 15)
0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1
DUTY CYCLE
FIGURE 7. NORMALIZED RMS INPUT CURRENT FOR x = 0.8
NORMALIZED INPUT RMS RIPPLE CURRENT
x = 1
x = 0.75
x = 0.50
x = 0.25
x = 0
0
0.05
0.10
0.15
0.20
0.25
0.30
0.35
0.40
0.45
0.50
0.55
0.60
ISL6269
12
FN9177.3
June 25, 2009
MOSFET Selection and Considerations
Typically, a MOSFET cannot tolerate even brief excursions
beyond their maximum drain to source voltage rating. The
MOSFETs used in the power stage of the converter should
have a maximum V
DS
rating that exceeds the sum of the
upper voltage tolerance of the input power source and the
voltage spike that occurs when the MOSFET switches off.
There are several power MOSFETs readily available that are
optimized for DC/DC converter applications. The preferred
high-side MOSFET emphasizes low switch charge so that
the device spends the least amount of time dissipating
power in the linear region. Unlike the low-side MOSFET
which has the drain-source voltage clamped by its body
diode during turn off, the high-side MOSFET turns off with
V
IN
-V
OUT
-V
L
across it. The preferred low-side MOSFET
emphasizes low r
DS(ON)
when fully saturated to minimize
conduction loss.
For the low-side MOSFET, (LS), the power loss can be
assumed to be conductive only and is written as:
For the high-side MOSFET, (HS), its conduction loss is
written as:
For the high-side MOSFET, its switching loss is written as:
Where:
-I
VALLEY
is the difference of the DC component of the
inductor current minus 1/2 of the inductor ripple current
-I
PEAK
is the sum of the DC component of the inductor
current plus 1/2 of the inductor ripple current
-t
ON
is the time required to drive the device into
saturation
-t
OFF
is the time required to drive the device into cut-off
Selecting The Bootstrap Capacitor
The selection of the bootstrap capacitor is written as:
Where:
-Q
g
is the total gate charge required to turn on the
high-side MOSFET
- V
BOOT
, is the maximum allowed voltage decay across
the boot capacitor each time the high-side MOSFET is
switched on
As an example, suppose the high-side MOSFET has a total
gate charge Q
g
, of 25nC at V
GS
= 5V, and a V
BOOT
of
200mV. The calculated bootstrap capacitance is 0.125µF; for
a comfortable margin select a capacitor that is double the
calculated capacitance, in this example 0.22µF will suffice.
Use an X7R or X5R ceramic capacitor.
Layout Considerations
As a general rule, power should be on the bottom layer of
the PCB and weak analog or logic signals are on the top
layer of the PCB. The ground-plane layer should be adjacent
to the top layer to provide shielding. The ground plane layer
should have an island located under the IC, the
compensation components, and the FSET components. The
island should be connected to the rest of the ground plane
layer at one point.
Signal Ground and Power Ground
The bottom of the ISL6269 QFN package is the signal
ground (GND) terminal for analog and logic signals of the IC.
Connect the GND pad of the ISL6269 to the island of ground
plane under the top layer using several vias, for a robust
thermal and electrical conduction path. Connect the input
capacitors, the output capacitors, and the source of the
lower MOSFETs to the power ground plane.
PGND (PIN 10)
This is the return path for the pull-down of the LG low-side
MOSFET gate driver. Ideally, PGND should be connected to
the source of the low-side MOSFET with a low-resistance,
low-inductance path.
VIN (PIN 1)
The VIN pin should be connected close to the drain of the
high-side MOSFET, using a low resistance and low
inductance path.
VCC (PIN 2)
For best performance, place the decoupling capacitor very
close to the VCC and GND pins.
PVCC (PIN 12)
For best performance, place the decoupling capacitor very
close to the PVCC and PGND pins, preferably on the same
side of the PCB as the ISL6269 IC.
(EQ. 16)
P
CON_LS
I
LOAD
2
r
DS ON_LS
1D
(EQ. 17)
P
CON_HS
I
LOAD
2
r
DS ON_HS
D=
(EQ. 18)
P
SW_HS
V
IN
I
VALLEY
t
ON
f
SW
2
-----------------------------------------------------------------
V
IN
I
PEAK
t
OFF
f
SW
2
-------------------------------------------------------------
+=
C
BOOT
Q
g
V
BOOT
------------------------
=
(EQ. 19)
INDUCTOR
VIAS TO
GROUND
PLANE
VIN
VOUT
PHASE
NODE
GND
OUTPUT
CAPACITORS
LOW-SIDE
MOSFETS
INPUT
CAPACITORS
SCHOTTKY
DIODE
HIGH-SIDE
MOSFETS
FIGURE 8. TYPICAL POWER COMPONENT PLACEMENT
ISL6269

ISL6269CRZ-T

Mfr. #:
Manufacturer:
Renesas / Intersil
Description:
Switching Controllers GPU CNTRLR 16LD 4X4
Lifecycle:
New from this manufacturer.
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