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MAX8598ETE+T

P1-P3P4-P6P7-P9P10-P12P13-P15P16-P18P19-P21P22-P24
MAX8597/MAX8598/MAX8599
Low-Dropout, Wide-Input-Voltage,
Step-Down Controllers
16 ______________________________________________________________________________________
Setting the Output Voltage
Fixed Output Voltage
The output voltage is set by a resistor-divider network
from the output to GND with FB at the center tap (R4
and R5 in Figure 4). Select R4 between 5kΩ and 15kΩ
and calculate R5 by:
R5 = R4 x [( V
OUT
/ V
FB
) - 1]
Live Adjustable Output Voltage (see Figure 1)
Using the uncommitted operational amplifier, the
MAX8597 can be configured such that the output volt-
age is adjustable using a voltage source (V
ADJ
). The
following parameters must be defined before starting
the design:
• The minimum desired output voltage, V
OUT_MIN
• The maximum desired output voltage, V
OUT_MAX
• The desired input that corresponds to the minimum
output voltage, V
ADJ_MIN
• The desired input that corresponds to the maximum
output voltage, V
ADJ_MAX
Select V
AOUT
(uncommitted operational-amplifier out-
put) between 0.05V and 3V and V
AOUT_MAX
higher
than V
AOUT_MIN
. Calculate the required AIN+ reference
(V
AIN+
) as:
V
AIN+
is set using a resistor-divider from REFOUT to
GND (R6 and R7). Select R7 to be approximately 50kΩ
as a starting point and then calculate R6 as:
R6 = R7 x [(2.5V / V
AIN+
) - 1]
Select R4 to be 100kΩ and calculate R5 as:
Select R9 between 5kΩ and 15kΩ, then calculate R8
and R10 as follows:
where V
FB
is the feedback regulation voltage (0.6V with
REFIN connected to AVL).
Additionally, to minimize error, R6 and R7 should be
chosen such that:
Inductor Selection
There are several parameters that must be examined
when determining which inductor is to be used: input
voltage, output voltage, load current, switching fre-
quency, and LIR. LIR is the ratio of inductor current rip-
ple to DC load current. A higher LIR value allows for a
smaller inductor but results in higher losses and higher
output ripple. A good compromise between size and
efficiency is a 30% LIR. Once all the parameters are
chosen, the inductor value is determined as follows:
where f
S
is the switching frequency. Choose a standard
value close to the calculated value. The exact inductor
value is not critical and can be adjusted in order to
make trade-offs among size, cost, and efficiency.
Lower inductor values minimize size and cost, but also
increase the output ripple and reduce the efficiency
due to higher peak currents. On the other hand, higher
inductor values increase efficiency, but eventually
resistive losses due to extra turns of wire exceed the
benefit gained from lower AC current levels. Find a low-
loss inductor having the lowest possible DC resistance
that fits the allotted dimensions. Ferrite cores are often
the best choice, although powdered iron is inexpensive
and can work well up to 300kHz. The chosen inductor’s
saturation current rating must exceed the peak inductor
current determined as:
Input Capacitor
The input filter capacitor reduces peak currents drawn
from the power source and reduces noise and voltage
ripple on the input caused by the circuit’s switching.
The input capacitor must meet the ripple current
requirement (I
RMS
) imposed by the switching currents
defined by the following equation:
I
IVVV
V
RMS
LOAD OUT IN OUT
IN
=
××
()
−
II
LIR
I
PEAK LOAD MAX LOAD MAX
=+
⎛
⎝
⎜
⎞
⎠
⎟
×
() ()
2
L
VxVV
V x f x I x LIR
OUT IN OUT
IN S LOAD MAX
()
=
()
−
RR
RR
RR
RR
67
67
45
45
×
+
=
×
+
R
RR V V
VR VV R
OUT MAX FB
FB FB AOUT MIN
10
89
89
_
_
=
××
()
×
()
+
()
×
[]
−
−
R
VVVV V VV V
VVVVV
R
OUT MIN FB FB AOUT MIN OUT MAX FB AOUT MAX FB
OUT MAX FB OUT MIN FB FB
8 9
____
__
=
()
×
()
+
()
×
()
[ ]
()()
()
×
×
−− − −
−− −
R
VV R
VV
AIN AOUT MIN
ADJ MAX AIN
5
4
( )
( )
_
_
=
×
+
+
−
−
V
VVVV
VV V V
AIN
AOUT MAX ADJ MAX AOUT MIN ADJ MIN
ADJ MAX ADJ MIN AOUT MAX AOUT MIN
+
=
××
+
−
−−
()( )
__ __
__ _ _
MAX8597/MAX8598/MAX8599
Low-Dropout, Wide-Input-Voltage,
Step-Down Controllers
______________________________________________________________________________________ 17
I
RMS
has a maximum value when the input voltage
equals twice the output voltage (V
IN
= 2 x V
OUT
), so
I
RMS(MAX)
= I
LOAD
/ 2. Ceramic capacitors are recom-
mended due to their low ESR and ESL at high frequen-
cy, with relatively lower cost. Choose a capacitor that
exhibits less than 10°C temperature rise at the maximum
operating RMS current for optimum long-term reliability.
Output Capacitor
The key selection parameters for the output capacitor
are the actual capacitance value, the equivalent series
resistance (ESR), the equivalent series inductance
(ESL), and the voltage-rating requirements, which
affect the overall stability, output ripple voltage, and
transient response. The output ripple has three compo-
nents: variations in the charge stored in the output
capacitor, voltage drop across the capacitor’s ESR,
and voltage drop across the capacitor’s ESL, caused
by the current into and out of the capacitor. The follow-
ing equations estimate the worst-case ripple:
where I
P-P
is the peak-to-peak inductor current.
The response to a load transient depends on the select-
ed output capacitor. After a load transient, the output
instantly changes by (ESR x ∆I
LOAD
) + (ESL x di/dt).
Before the controller can respond, the output deviates
further depending on the inductor and output capacitor
values. After a short period of time (see the Typical
Operating Characteristics), the controller responds by
regulating the output voltage back to its nominal state.
The controller response time depends on the closed-
loop bandwidth. With higher bandwidth, the response
time is faster, preventing the output capacitor voltage
from further deviation from its regulation value. Do not
exceed the capacitor’s voltage or ripple current ratings.
MOSFET Selection
The MAX8597/MAX8598/MAX8599 controllers drive
external, logic-level, n-channel MOSFETs as the circuit-
switch elements. The key selection parameters are:
• On-resistance (R
DS(ON)
): the lower, the better.
• Maximum drain-to-source voltage (V
DSS
): should be
at least 20% higher than the input supply rail at the
high-side MOSFET’s drain.
• Gate charges (Q
g
, Q
gd
, Q
gs
): the lower, the better.
Choose MOSFETs with R
DS(ON)
rated at V
GS
= 4.5V. For
a good compromise between efficiency and cost,
choose the high-side MOSFET that has conduction loss
equal to the switching loss at the nominal input voltage
and maximum output current. For the low-side MOSFET,
make sure it does not spuriously turn on due to dv/dt
caused by the high-side MOSFET turning on, resulting in
efficiency degrading shoot-through current. MOSFETs
with a lower Q
gd
/Q
gs
ratio have higher immunity to dv/dt.
For proper thermal-management design, the power dis-
sipation must be calculated at the desired maximum
operating junction temperature, maximum output current,
and worst-case input voltage (for low-side MOSFET,
worst case is at V
IN(MAX)
; for high-side MOSFET, it could
be either at V
IN(MIN)
or V
IN(MAX)
).
High-side and low-side MOSFETs have different loss
components due to the circuit operation. The low-side
MOSFET operates as a zero-voltage switch; therefore,
the major losses are the channel-conduction loss
(P
LSCC
) and the body-diode conduction loss (P
LSDC
):
P
LSCC
= [1 - (V
OUT
/ V
IN
)] x (I
LOAD
)
2
x R
DS(ON)
P
LSDC
= 2 x I
LOAD
x V
F
x t
dt
x f
S
where V
F
is the body-diode forward-voltage drop, t
dt
is
the dead-time between the high-side MOSFET and the
low-side MOSFET switching transitions, and f
S
is the
switching frequency. The high-side MOSFET operates
as a duty-cycle control switch and has the following
major losses: the channel-conduction loss (P
HSCC
), the
V-I overlapping switching loss (P
HSSW
), and the drive
loss (P
HSDR
). The high-side MOSFET does not have
body-diode conduction loss because the diode never
conducts current:
P
HSCC
= (V
OUT
/ V
IN
) x I
LOAD
2
x R
DS(ON)
Use R
DS(ON)
at T
J(MAX)
:
P
HSSW
= V
IN
x I
LOAD
x f
S
x [(Q
gs
+ Q
gd
) / I
GATE
]
where I
GATE
is the average DH-high driver output-cur-
rent capability determined by:
I
GATE
= 2.5 / (R
DH
+ R
GATE
)
VV V V
V I ESR
V
V ESL
L ESL
V
I
Cf
I
VV
fL
V
V
RIPPLE RIPPLE ESR RIPPLE ESL RIPPLE C
RIPPLE ESR P P
RIPPLE ESL
IN
RIPPLE C
PP
OUT S
PP
IN OUT
S
OUT
IN
=++
=×
=
×
+
=
××
=
×
⎛
⎝
⎜
⎞
⎠
⎟
×
⎛
⎝
⎜
⎞
⎠
⎟
−
−
−
−
() () ()
()
()
()
8
MAX8597/MAX8598/MAX8599
where R
DH
is the high-side MOSFET driver’s average
on-resistance (1.25Ω typ) and R
GATE
is the internal
gate resistance of the MOSFET (typically 0.5Ω to 2Ω):
P
HSDR
= Q
gs
x V
GS
x f
S
x R
GATE
/ (R
GATE
+ R
DH
)
where V
GS
~ V
VL
= 5V.
In addition to the losses above, add approximately
20% more for additional losses due to MOSFET output
capacitances and low-side MOSFET body-diode
reverse-recovery charge dissipated in the high-side
MOSFET that exists, but is not well defined in the
MOSFET data sheet. Refer to the MOSFET data sheet
for thermal-resistance specification to calculate the
PC board area needed to maintain the desired maxi-
mum operating junction temperature with the above-
calculated power dissipation. To reduce EMI caused
by switching noise, add a 0.1µF or larger ceramic
capacitor from the high-side switch drain to the low-
side switch source or add resistors in series with DH
and DL to slow down the switching transitions.
However, adding a series resistor increases the power
dissipation of the MOSFETs, so be sure this does not
overheat the MOSFETs. The minimum load current must
exceed the high-side MOSFET’s maximum leakage plus
the maximum LX bias current over temperature.
Setting the Current-Limit
The MAX8597/MAX8598/MAX8599 controllers sense
the peak inductor current to provide constant-current
and hiccup current limit. The peak current-limit thresh-
old is set by an external resistor (R2 in Figure 1) togeth-
er with the internal current sink of 200µA. The voltage
drop across the resistor R2 due to the 200µA current
sets the maximum peak inductor current that can flow
through the high-side MOSFET or the optional current-
sense resistor (between the high-side MOSFET source
and LX) by the equations below:
I
PEAK(MAX)
= 200µA x R2 / R
DSON(HSFET)
I
PEAK(MAX)
= 200µA x R2 / R
SENSE
The actual corresponding maximum load current is
lower than the I
PEAK(MAX)
by half of the inductor ripple
current. If the R
DS(ON)
of the high-side MOSFET is used
for current sensing, use the maximum R
DS(ON)
at the
highest operating junction temperature to avoid false
tripping of the current limit at elevated temperature.
Consideration should also be given to the tolerance of
the 200µA current sink. When the R
DS(ON)
of the high-
side MOSFET is used for current sensing, ringing on
the LX voltage waveform can interfere with the current
limit. Below is the procedure for selecting the value of the
series RC snubber circuit (R14 and C14 in Figure 1):
1) Connect a scope probe to measure V
LX
to GND,
and observe the ringing frequency, f
R
.
2) Find the capacitor value (connected from LX to
GND) that reduces the ringing frequency by half.
The circuit parasitic capacitance (C
PAR
) at LX is
then equal to 1/3 the value of the added capaci-
tance above. The circuit parasitic inductance (L
PAR
)
is calculated by:
The resistor for critical dampening (R14) is equal to 2π x
f
R
x L
PAR
. Adjust the resistor value up or down to tailor
the desired damping and the peak voltage excursion.
The capacitor (C14) should be at least 2 to 4 times the
value of the C
PAR
in order to be effective. The power
loss of the snubber circuit is dissipated in the resistor
(R14) and is calculated as:
P
R14
= C14 x (V
IN
)
2
x f
S
where V
IN
is the input voltage and f
S
is the switching
frequency. Choose an R14 power rating that meets the
specific application’s derating rule for the power dissi-
pation calculated.
Additionally, there is parasitic inductance of the cur-
rent-sensing element, whether the high-side MOSFET
(L
SENSE_FET
) or the optional current-sense resistor
(L
RSENSE
) are used, which is in series with the output
filter inductor. This parasitic inductance, together with
the output inductor, forms an inductive divider and
causes error in the current-sensing voltage. To com-
pensate for this error, a series RC circuit can be added
in parallel with the sensing element (see Figure 5). The
RC time constant should equal L
RSENSE
/ R
SENSE
, or
L
SENSE_FET
/ R
DS(ON)
. First, set the value of R equal to or
less than R2 / 100. Then, the value of C is calculated as:
C = L
RSENSE
/ (R
SENSE
x R) or
C = L
SENSE_FET
/ (R
DS(ON)
x R)
Any PC board trace inductance in series with the sens-
ing element and output inductor should be added to
the specified FET or resistor inductance per the
respective manufacturer’s data sheet. For the case of
L
fC
PAR
R PAR
=
×
()
×
1
2
2
π
Low-Dropout, Wide-Input-Voltage,
Step-Down Controllers
18 ______________________________________________________________________________________
P1-P3P4-P6P7-P9P10-P12P13-P15P16-P18P19-P21P22-P24

MAX8598ETE+T

Mfr. #:
Buy MAX8598ETE+T
Manufacturer:
Maxim Integrated
Description:
Switching Controllers Step-Down Controller
Lifecycle:
New from this manufacturer.
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