AD8314
Rev. B | Page 14 of 20
When connected in a PA control loop, as shown in Figure 34,
the voltage V
UP
is not explicitly used but is implicated in again
setting up the required averaging time, by choice of C
F
.
However, now the effective loop response time is a much more
complicated function of the PA’s gain-control characteristics,
which are very nonlinear. A complete solution requires specific
knowledge of the power amplifier.
The transient response of this control loop is determined by the
filter capacitor, C
F.
When this is large, the loop is unconditionally
stable (by virtue of the dominant pole generated by this
capacitor), but the response is sluggish. The minimum value
ensuring stability should be used, requiring full attention to the
particulars of the power amplifier control function. Because this
is invariably nonlinear, the choice must be made for the worst-
case condition, which usually corresponds to the smallest
output from the PA, where the gain function is steepest. In
practice, an improvement in loop dynamics can often be
achieved by adding a response zero, formed by a resistor in
series with C
F
.
POWER-ON AND ENABLE GLITCH
As previously mentioned, the AD8314 can be put into a low
power mode by pulling the ENBL pin to ground. This reduces
the quiescent current from 4.5 mA to 20 µA. Alternatively, the
supply can be turned off to eliminate the quiescent current.
Figure 16 and Figure 26 show the behavior of the V_DN output
under these two conditions (in
Figure 26, ENBL is tied to
VPOS). The glitch that results in both cases can be reduced by
loading the V_DN output.
INPUT COUPLING OPTIONS
The internal 5 pF coupling capacitor of the AD8314, along with
the low frequency input impedance of 3 kΩ, gives a high-pass
input corner frequency of approximately 16 MHz. This sets the
minimum operating frequency.
Figure 35 through Figure 37
show three options for input coupling. A broadband resistive
match can be implemented by connecting a shunt resistor to
ground at RFIN (see
Figure 35). This 52.3 Ω resistor (other
values can also be used to select different overall input
impedances) combines with the input impedance of the
AD8314 (3 kΩ||2 pF) to give a broadband input impedance of
50 Ω. While the input resistance and capacitance (CIN and
RIN) varies by approximately ±20% from device to device, the
dominance of the external shunt resistor means that the variation
in the overall input impedance is close to the tolerance of the
external resistor.
At frequencies above 2 GHz, the input impedance drops below
250 Ω (see
Figure 12), so it is appropriate to use a larger value
shunt resistor. This value is calculated by plotting the input
impedance (resistance and capacitance) on a Smith Chart and
choosing the best value shunt resistor to bring the input
impedance closest to the center of the chart. At 2.5 GHz, a
shunt resistor of 165 Ω is recommended.
A reactive match can also be implemented as shown in
Figure 36.
This is not recommended at low frequencies as device
tolerances dramatically varies the quality of the match because
of the large input resistance. For low frequencies,
Figure 35 or
Figure 37 is recommended.
In
Figure 36, the matching components are drawn as general
reactances. Depending on the frequency, the input impedance at
that frequency, and the availability of standard value components,
either a capacitor or an inductor is used. As in the previous
case, the input impedance at a particular frequency is plotted on
a Smith Chart and matching components are chosen (shunt or
Series L, shunt or Series C) to move the impedance to the center
of the chart.
Table 5 gives standard component values for some
popular frequencies. Matching components for other frequencies
can be calculated using the input resistance and reactance data
over frequency, which is given in
Figure 12. Note that the
reactance is plotted as though it appears in parallel with the
input impedance (which it does because the reactance is
primarily due to input capacitance).
The impedance matching characteristics of a reactive matching
network provide voltage gain ahead of the AD8314; this increases
the device sensitivity (see
Table 5). The voltage gain is calculated by
1
2
log20
10
R
R
GainVoltage
dB
=
where R2 is the input impedance of the AD8314, and R1 is the
source impedance to which the AD8314 is being matched. Note
that this gain is only achieved for a perfect match. Component
tolerances and the use of standard values tend to reduce gain.
50Ω SOURCE
R
SHUNT
52.3Ω
50Ω
C
IN
R
IN
C
C
AD8314
RFIN
V
BIAS
01086-035
Figure 35. Broadband Resistive
50Ω SOURCE
X2
X1
50Ω
C
IN
R
IN
C
C
AD8314
RFIN
V
BIAS
01086-036
Figure 36. Narrowband Reactive
STRIPLINE
C
IN
R
IN
C
C
AD8314
RFIN
V
BIAS
50Ω
R
ATTN
01086-037
Figure 37. Series Attenuation
