NCP1216AFORWGEVB
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3
Description of Converter Connection
Capacitors C1, C2, C3 and inductor L1 form the input
filter. Diode D3, capacitor C5 and resistor R5 provide the
primary clamping network which combats leakage
inductance between the reset winding and the primary
winding. The link between both windings occurs via D2
when the switch is off. Transformer T2 with diode D1 and
resistors R2, R3 serve as the primary current sensing circuit.
Thanks to low insertion losses, the final efficiency of the
converter benefits greatly from this configuration. IC1 is the
main driving circuit of the power converter. The secondary
circuitry has D4A as the forward diode and D4B as the
freewheeling diode. Capacitor C6 offers a path for
common-mode (CM) currents circulating via the various
transformer stray capacitances during switching events.
Resistors R7, R8, R9, and R10 together with capacitor C12,
shunt regulator IC3, and optocoupler IC2 form an isolated
feedback circuit for output voltage regulation. A snubber
network (R6, C7) is connected across inductor L2 in order
to damp high frequency oscillations. L2, C8, C9 and C10
form the basic LC output filter. L3 and C11 form an
additional output filter to reduce high frequency noise.
Design considerations for various sections of the
converter are described below.
Transformer Design
In a forward converter, the core magnetization is ensured
by applying a voltage V
in
on the primary side. This action
creates the core flux f which links both primary and
secondary windings. Using Faraday’s law, we can write that
E = N.df/dt, where E is the voltage generated by a winding
of N turns, energized by a flux f. By integrating this formula,
and rearranging it in terms of the input voltage V
in
and the
on time ton, we can see that the internal flux depends on the
volt-second product:
V
in
@ t
on
+ N @ f + N @ A
e
@ B
(eq. 1)
where:
A
e
is the total core area
B is the core flux density
Thus, the maximum core flux density DB
MAX
and the peak
primary magnetization current I
PKMAG
of the transformer
are given by the primary inductance value L1 and the
maximum input voltage according to equations (2) and (3):
I
PKMAG
+
V
in max
L
1
@
1
f
op
@ d
max
(eq. 2)
DB
MAX
+
V
in max
@ d
max
N
p
@ f
op
@ A
e
(eq. 3)
where:
V
in
max
is the maximum input voltage
L
1
is the primary winding inductance
f
op
is the operating frequency
ą
d
max
is the maximum duty cycle
N
p
is the count of the primary turns
The primary magnetization current does not directly
participate in the energy transfer and cause additive losses
on the power switch and the primary winding. When the
switch is off, the transformer core must be reset in order to
let the internal flux return to zero. This is done via a
dedicated reset circuit. Consequently the magnetizing
current Imag must be kept smaller than the productive
component of the primary current.
The core flux density excursion DB has to be chosen with
respect to the characteristics of the core material: the
saturation flux density Bmax or Bsat, the residual flux
density Br, hysteretic losses and the core temperature
behavior. With respect to these characteristics, the flux
density excursion in high frequency converters should be
between 0.15 T and 0.2 T. If a higher value is chosen, greater
losses will be generated. The primary turn count N
p
can be
calculated by rearranging equation 4:
N
p
+
V
in max
@ d
max
DB
MAX
@ f
op
@ A
e
(eq. 4)
For an EFD25 core with a total core area of 58 mm
2
(DB
max
= 0.2 T, V
in
max
= 80 V, f
op
= 100 kHz and
maximum duty cycle d
max
= 0.5) then the number of
primary turns N
p
= 35.
The number of reset winding turns depends on design
tradeoffs. When the number of turns of the reset winding is
lower than the that of the primary winding, the reflected
voltage on the power switch drain will be lower than
2 V
in max
. However, this limits the maximum duty cycle
excursion to less than 50%. Conversely, if the reset turns are
larger than the primary turns, the maximum allowed duty
cycle will increase but the MOSFET voltage stress will
exceed 2 V
in
max
. Due to these issues, the practical number
of turns for the reset winding is usually chosen to be the same
as the primary winding, or a 1:1 ratio. It is important to
provide a very good coupling between these two windings.
A high leakage inductance between these windings would
require a hard voltage clamp that would hurt the converter
efficiency.
The number of turns on the secondary winding N
s
can be
obtained from equation 5:
N
s
+ N
p
@
V
out
d
max
) V
f
V
in min
(eq. 5)
where:
V
out
is the desired output voltage
V
f
is the voltage drop of the output rectifier
V
in
min
is the minimum input voltage
In the example using the EFD 25, equation (5) gives
N
s
= 25 turns.
The primary and the secondary windings must be wound
to limit the skin effect. This can be done by using several
wires wound in parallel. The maximum diameter D
max
(in mm) of each single wire in the winding is given by
equation 6:
