OMO ElectronicOMO Electronic
Expand menu
Hello, Sign inMy Account
0 Cart

LTC3737EUF#TRPBF

P1-P3P4-P6P7-P9P10-P12P13-P15P16-P18P19-P21P22-P24
13
LTC3737
3737fa
APPLICATIO S I FOR ATIO
WUUU
about 10°C in most applications. For a maximum ambient
temperature of 70°C, using ρ
80°C
~ 1.3 in the above
equation is a reasonable choice.
The power dissipated in the MOSFET strongly depends on
its respective duty cycles and load current. When the
LTC3737 is operating in continuous mode, the duty cycles
for the MOSFET are:
Duty Cycle =
V
OUT
+
+
V
VV
D
IN D
The MOSFET power dissipations at maximum output
current are:
P
P
=
+
+
()
+
VV
VV
IR
VI C f
OUT D
IN D
OUT MAX T DS ON
IN OUT MAX RSS OSC
••••
•••
() ()
()
2
2
2r
The MOSFET has I
2
R losses and the P
P
equation includes
an additional term for transition losses, which are largest
at high input voltages.
Using a Sense Resistor
A sense resistor R
SENSE
can be connected between SENSE
+
and SW to sense the output load current. In this case, the
source of the P-channel MOSFET is connected to the SW
pin and the drain is not connected to any pin of the
LTC3737. Therefore, the current comparator monitors the
voltage developed across R
SENSE
instead of V
DS
of the
P-channel MOSFET. The output current that the LTC3737
can provide in this case is given by:
I
V
R
I
OUT MAX
SENSE MAX
SENSE
RIPPLE
()
()
–=
∆
2
Setting ripple current as 40% of I
OUT(MAX)
and using
Figure 2 to choose SF, the value of R
SENSE
is:
RSF
V
I
SENSE
SENSE MAX
OUT MAX
=
∆
5
6
••
()
()
(See the R
DS(ON)
selection in Power MOSFET Selection).
Variation in the resistance of a sense resistor is much
smaller than the variation in on-resistance of the external
MOSFET. Therefore the load current is well controlled, and
the system is more stable with a sense resistor. However
the sense resistor causes extra I
2
R losses in addition to the
I
2
R losses of the MOSFET. Therefore, using a sense
resistor lowers the efficiency of LTC3737, especially for
large load current.
Operating Frequency and Synchronization
The choice of operating frequency, f
OSC
, is a tradeoff
between efficiency and component size. Low frequency
operation improves efficiency by reducing MOSFET switch-
ing losses, both gate charge loss and transition loss.
However, lower frequency operation requires more induc-
tance for a given amount of ripple current.
The internal oscillator for each of the LTC3737’s control-
lers runs at a nominal 550kHz frequency when the PLLLPF
pin is left floating and the SYNC/MODE pin is a DC low or
high. Pulling the PLLLPF to V
IN
selects 750kHz operation;
pulling the PLLLPF to GND selects 300kHz operation.
Alternatively, the LTC3737 will phase lock to a clock signal
applied to the SYNC/MODE pin with a frequency between
250kHz and 850kHz (see Phase-Locked Loop and Fre-
quency Synchronization).
Inductor Value Calculation
Given the desired input and output voltages, the inductor
value and operating frequency, f
OSC
,
directly determine
the inductor’s peak-to-peak ripple current:
IVV
VVVV
fL
RIPPLE IN OUT
OUT D IN D
OSC
=
()
+
()
+
()
⎛
⎝
⎜
⎞
⎠
⎟
–
/
•
Lower ripple current reduces core losses in the inductor,
ESR losses in the output capacitors, and output voltage
ripple. Thus, highest efficiency operation is obtained at
low frequency with a small ripple current. Achieving this,
however, requires a large inductor.
A reasonable starting point is to choose a ripple current
that is about 40% of I
OUT(MAX).
Note that the largest ripple
current occurs at the highest input voltage. To guarantee
14
LTC3737
3737fa
APPLICATIO S I FOR ATIO
WUUU
Kool Mµ is a registered trademark of Magnetics, Inc.
that ripple current does not exceed a specified maximum,
the inductor should be chosen according to:
L
VV
fI
VV
VV
IN OUT
OSC RIPPLE
OUT D
IN D
≥
+
+
–
•
•
Burst Mode Operation Considerations
The choice of R
DS(ON)
and inductor value also determines
the load current at which the LTC3737 enters Burst Mode
operation. When bursting, the controller clamps the peak
inductor current to approximately:
I
V
R
BURST PEAK
SENSE MAX
DS ON
()
()
()
•=
∆
1
4
The corresponding average current depends on the amount
of ripple current. Lower inductor values (higher I
RIPPLE
)
will reduce the load current at which Burst Mode operation
begins.
The ripple current is normally set so that the inductor
current is continuous during the burst periods. Therefore,
I
RIPPLE
≤ I
BURST(PEAK)
This implies a minimum inductance of:
L
VV
fI
VV
VV
MIN
IN OUT
OSC BURST PEAK
OUT D
IN D
≥
+
+
–
•
•
()
A smaller value than L
MIN
could be used in the circuit,
although the inductor current will not be continuous
during burst periods, which will result in slightly lower
efficiency. In general, though, it is a good idea to keep
I
RIPPLE
comparable to I
BURST(PEAK)
.
Inductor Core Selection
Once the value of L is known, the type of inductor must be
selected. High efficiency converters generally cannot af-
ford the core loss found in low cost powdered iron cores,
forcing the use of more expensive ferrite, molypermalloy
or Kool Mµ
®
cores. Actual core loss is independent of core
size for a fixed inductor value, but is very dependent on the
inductance selected. As inductance increases, core losses
go down. Unfortunately, increased inductance requires
more turns of wire and therefore copper losses will in-
crease. Ferrite designs have very low core losses and are
preferred at high switching frequencies, so design goals
can concentrate on copper loss and preventing saturation.
Ferrite core material saturates “hard”, which means that
inductance collapses abruptly when the peak design cur-
rent is exceeded. Core saturation results in an abrupt
increase in inductor ripple current and consequent output
voltage ripple. Do not allow the core to saturate!
Molypermalloy (from Magnetics, Inc.) is a very good, low
loss core material for toroids, but is more expensive than
ferrite. A reasonable compromise from the same manu-
facturer is Kool Mµ. Toroids are very space efficient,
especially when several layers of wire can be used, while
inductors wound on bobbins are generally easier to sur-
face mount. However, designs for surface mount that do
not increase the height significantly are available from
Coiltronics, Coilcraft, Dale and Sumida.
Output Diode Selection
The catch diode carries load current during the switch off
time of the power MOSFETs . The average diode current is
therefore dependent on the P-channel MOSFET duty cycle.
At high input voltages, the diode conducts most of the
time. As V
IN
approaches V
OUT
, the diode conducts for only
a small fraction of the time. The most stressful condition
for the diode is when the output is short circuited. Under
this condition, the diode must safely handle I
PEAK
at close
to 100% duty cycle. Therefore, it is important to ad-
equately specify the diode peak current and average power
dissipation so as not to exceed the diode’s ratings.
Under normal conditions, the average current conducted
by the diode is:
I
VV
VV
I
D
IN OUT
IN D
OUT
=
+
–
•
The allowable forward voltage drop in the diode is calcu-
lated from the maximum short-circuit current as:
V
P
I
F
D
PEAK
≈
15
LTC3737
3737fa
APPLICATIO S I FOR ATIO
WUUU
where P
D
is the allowable power dissipation and will be
determined by efficiency and/or thermal requirements.
A Schottky diode is a good choice for low forward drop and
fast switching time. Remember to keep lead length short
and observe proper grounding to avoid ringing and
increased dissipation.
C
IN
and C
OUT
Selection
The selection of C
IN
is simplified by the 2-phase architec-
ture and its impact on the worst-case RMS current drawn
through the input network (battery/fuse/capacitor). It can
be shown that the worst-case capacitor RMS current
occurs when only one controller is operating. The control-
ler with the highest V
OUT
• I
OUT
product needs to be used
in the formula below to determine the maximum RMS
capacitor current requirement. Increasing the output cur-
rent drawn from the other controller will actually decrease
the input RMS ripple current from its maximum value. The
out-of-phase technique typically reduces the input
capacitor’s RMS ripple current by a factor of 30% to 70%
when compared to a single phase power supply solution.
In continuous mode, the source current of the P-channel
MOSFET is a square wave of duty cycle (V
OUT
+ V
D
)/
(V
IN
+ V
D
). To prevent large voltage transients, a low ESR
capacitor sized for the maximum RMS current of one
channel must be used. The maximum RMS capacitor
current is given by:
C
I
VV
VVVV
IN
MAX
IN D
OUT D IN OUT
Required I
RMS
≈
+
+
()( )
[]
–
/12
This formula has a maximum at V
IN
= 2V
OUT
+ V
D
, where
I
RMS
= I
OUT
/2. This simple worst-case condition is com-
monly used for design because even significant deviations
do not offer much relief. Note that capacitor manufactur-
ers’ ripple current ratings are often based on only 2000
hours of life. This makes it advisable to further derate the
capacitor or to choose a capacitor rated at a higher
temperature than required. Several capacitors may be
paralleled to meet size or height requirements in the
design. Due to the high operating frequency of the LTC3737,
ceramic capacitors can also be used for C
IN
. Always
consult the manufacturer if there is any question.
The benefit of the LTC3737 2-phase operation can be
calculated by using the equation above for the higher
power controller and then calculating the loss that would
have resulted if both controller channels switched on at
the same time. The total RMS power lost is lower when
both controllers are operating due to the reduced overlap
of current pulses required through the input capacitor’s
ESR. This is why the input capacitor’s requirement calcu-
lated above for the worst-case controller is adequate for
the dual controller design. Also, the input protection fuse
resistance, battery resistance, and PC board trace resis-
tance losses are also reduced due to the reduced peak
currents in a 2-phase system. The overall benefit of a
multiphase design will only be fully realized when the
source impedance of the power supply/battery is included
in the efficiency testing. The sources of the P-channel
MOSFETs should be placed within 1cm of each other and
share a common C
IN
(s). Separating the sourced and C
IN
may produce undesirable voltage and current resonances
at V
IN
.
A small (0.1µF to 1µF) bypass capacitor between the chip
V
IN
pin and ground, placed close to the LTC3737, is also
suggested. A 10Ω resistor placed between C
IN
and the V
IN
pin provides further isolation between the two channels.
The selection of C
OUT
is driven by the effective series
resistance (ESR). Typically, once the ESR requirement is
satisfied, the capacitance is adequate for filtering. The
output ripple (∆V
OUT
) is approximated by:
∆≈ +
⎛
⎝
⎜
⎞
⎠
⎟
V I ESR
fC
OUT RIPPLE
OUT
1
8
where f is the operating frequency, C
OUT
is the output
capacitance and I
RIPPLE
is the ripple current in the induc-
tor. The output ripple is highest at maximum input voltage
since I
RIPPLE
increases with input voltage.
P1-P3P4-P6P7-P9P10-P12P13-P15P16-P18P19-P21P22-P24

LTC3737EUF#TRPBF

Mfr. #:
Buy LTC3737EUF#TRPBF
Manufacturer:
Analog Devices / Linear Technology
Description:
Switching Voltage Regulators 2-Phase Controller w/Tracking
Lifecycle:
New from this manufacturer.
Delivery:
DHL FedEx Ups TNT EMS
Payment:
T/T Paypal Visa MoneyGram Western Union

Products related to this Datasheet