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LT1766EGN-5#PBF

P1-P3P4-P6P7-P9P10-P12P13-P15P16-P18P19-P21P22-P24P25-P27P28-P30
LT1766/LT1766-5
10
1766fc
APPLICATIONS INFORMATION
FEEDBACK PIN FUNCTIONS
The feedback (FB) pin on the LT1766 is used to set output
voltage and provide several overload protection features.
The fi rst part of this section deals with selecting resistors
to set output voltage and the remaining part talks about
foldback frequency and current limiting created by the FB
pin. Please read both parts before committing to a fi nal
design. The 5V fi xed output voltage part (LT1766-5) has
internal divider resistors and the FB pin is renamed SENSE,
connected directly to the output.
The suggested value for the output divider resistor (see
Figure 2) from FB to ground (R2) is 5k or less, and a
formula for R1 is shown below. The output voltage error
caused by ignoring the input bias current on the FB pin
is less than 0.25% with R2 = 5k. A table of standard 1%
values is shown in Table 1 for common output voltages.
Please read the following if divider resistors are increased
above the suggested values.
R
RV
OUT
1
2122
122
=
−
()
.
.
Table 1
OUTPUT
VOLTAGE
(V)
R2
(kΩ)
R1
(NEAREST 1%)
(kΩ)
% ERROR AT OUTPUT
DUE TO DISCREET 1%
RESISTOR STEPS
34.997.32 +0.32
3.3 4.99 8.45 –0.43
54.9915.4 –0.30
64.7518.7 +0.38
84.4724.9 +0.20
10 4.32 30.9 –0.54
12 4.12 36.5 +0.24
15 4.12 46.4 –0.27
More Than Just Voltage Feedback
The feedback pin is used for more than just output voltage
sensing. It also reduces switching frequency and current
limit when output voltage is very low (see the Frequency
Foldback graph in Typical Performance Characteristics).
This is done to control power dissipation in both the IC
and in the external diode and inductor during short-cir-
cuit conditions. A shorted output requires the switching
regulator to operate at very low duty cycles, and the
average current through the diode and inductor is equal
to the short-circuit current limit of the switch (typically 2A
for the LT1766, folding back to less than 1A). Minimum
switch on-time limitations would prevent the switcher
from attaining a suffi ciently low duty cycle if switching
frequency were maintained at 200kHz, so frequency is
reduced by about 5:1 when the feedback pin voltage drops
below 0.8V (see Frequency Foldback graph). This does
not affect operation with normal load conditions; one
simply sees a gear shift in switching frequency during
start-up as the output voltage rises.
In addition to lower switching frequency, the LT1766 also
operates at lower switch current limit when the feedback
pin voltage drops below 0.6V. Q2 in Figure 2 performs
this function by clamping the V
C
pin to a voltage less than
its normal 2.1V upper clamp level. This
foldback current
limit
greatly reduces power dissipation in the IC, diode
and inductor during short-circuit conditions. External syn-
chronization is also disabled to prevent interference with
foldback operation. Again, it is nearly transparent to the user
under normal load conditions. The only loads that may be
affected are current source loads which maintain full load
current with output voltage less than 50% of fi nal value.
In these rare situations the feedback pin can be clamped
above 0.6V with an external diode to defeat foldback cur-
rent limit.
Caution:
clamping the feedback pin means that
frequency shifting will also be defeated, so a combination
of high input voltage and dead shorted output may cause
the LT1766 to lose control of current limit.
The internal circuitry which forces reduced switching
frequency also causes current to fl ow out of the feedback
pin when output voltage is low. The equivalent circuitry is
shown in Figure 2. Q1 is completely off during normal op-
eration. If the FB pin falls below 0.8V, Q1 begins to conduct
current and reduces frequency at the rate of approximately
1.4kHz/μA. To ensure adequate frequency foldback (under
worst-case short-circuit conditions), the external divider
Thevinin resistance must be low enough to pull 115μA out
of the FB pin with 0.44V on the pin (R
DIV
≤ 3.8k).
The net
result is that reductions in frequency and current limit are
affected by output voltage divider impedance. Although
LT1766/LT1766-5
11
1766fc
APPLICATIONS INFORMATION
divider impedance is not critical, caution should be used if
resistors are increased beyond the suggested values and
short-circuit conditions occur with high input voltage.
High
frequency pickup will increase and the protection accorded
by frequency and current foldback will decrease.
CHOOSING THE INDUCTOR
For most applications, the output inductor will fall into
the range of 15μH to 100μH. Lower values are chosen to
reduce physical size of the inductor. Higher values allow
more output current because they reduce peak current
seen by the LT1766 switch, which has a 1.5A limit. Higher
values also reduce output ripple voltage.
When choosing an inductor you will need to consider
output ripple voltage, maximum load current, peak induc-
tor current and fault current in the inductor. In addition,
other factors such as core and copper losses, allowable
component height, EMI, saturation and cost should also
be considered. The following procedure is suggested
as a way of handling these somewhat complicated and
confl icting requirements.
Output Ripple Voltage
Figure 3 shows a typical output ripple voltage wave-
form for the LT1766. Ripple voltage is determined by
ripple current (I
LP-P
) through the inductor and the high
frequency impedance of the output capacitor. The fol-
lowing equations will help in choosing the required
inductor value to achieve a desirable output ripple volt-
age level. If output ripple voltage is of less importance,
the subsequent suggestions in Peak Inductor and Fault
Current and EMI will additionally help in the selection of
the inductor value.
Peak-to-peak output ripple voltage is the sum of a triwave
(created by peak-to-peak ripple current (I
LP-P
) times ESR)
and a square wave (created by parasitic inductance (ESL)
and ripple current slew rate). Capacitive reactance is as-
sumed to be small compared to ESR or ESL.
V I ESR ESL
dI
dt
RIPPLE LP P
=
()()
+
()
-
Figure 2. Frequency and Current Limit Foldback
–
+
1.2V
BUFFER
V
SW
L1
V
C
GND
TO SYNC CIRCUIT
1766 F02
TO FREQUENCY
SHIFTING
R3
1k
R4
2k
R1
C1
R2
5k
OUTPUT
5V
ERROR
AMPLIFIER
FB
1.4V
Q1
LT1766
Q2
+
Figure 3. LT1766 Ripple Voltage Waveform
2.5μs/DIV
40mV/DIV
V
OUT
AT I
OUT
= 1A
V
OUT
AT I
OUT
= 0.1A
INDUCTOR CURRENT
AT I
OUT
= 1A
INDUCTOR CURRENT
AT I
OUT
= 0.1A
0.5A/DIV
V
IN
= 40V
V
OUT
= 5V
L = 47μH
C = 100μF, 10V, 0.1Ω
1766 F03
LT1766/LT1766-5
12
1766fc
If maximum load current is 0.5A, for instance, a 0.5A
inductor may not survive a continuous 2A overload con-
dition. Dead shorts will actually be more gentle on the
inductor because the LT1766 has frequency and current
limit foldback.
Peak switch and inductor current can be signifi cantly higher
than output current, especially with smaller inductors
and lighter loads, so don’t omit this step. Powdered iron
cores are forgiving because they saturate softly, whereas
ferrite cores saturate abruptly. Other core materials fall
somewhere in between. The following formula assumes
continuous mode of operation, but errs only slightly on
where:
ESR = equivalent series resistance of the output
capacitor
ESL = equivalent series inductance of the output
capacitor
dI/dt = slew rate of inductor ripple current = V
IN
/L
Peak-to-peak ripple current (I
LP-P
) through the inductor
and into the output capacitor is typically chosen to be
between 20% and 40% of the maximum load current. It
is approximated by:
I
VVV
VfL
LP P
OUT IN OUT
IN
-
=
()( )
()()()
–
Example: with V
IN
= 40V, V
OUT
= 5V, L = 47μH, ESR = 0.1Ω
and ESL = 10nH, output ripple voltage can be approximated
as follows:
IA
dI
dt
VA
mV
RIPPLE
P-P
P-P
=
()
−
()
()
()()
=
==
=
()()
+
()()
()
=+=
−
−
−
540 5
40 47 10 200 10
0 465
40
47 10
10 0 85
0 465 0 1 10 10 10 0 85
0 0465 0 0085 55
63
6
6
96
••
.
•
•.
..• .
..
To reduce output ripple voltage further requires an increase
in the inductor value or a reduction in the capacitor ESR.
The latter can effect loop stability since the ESR forms
a useful zero in the overall loop response. Typically the
inductor value is adjusted with the trade-off being a
physically larger inductor with the possibility of increased
component height and cost. Choosing a smaller inductor
with lighter loads may result in discontinuous operation
but the LT1766 is designed to work well in both continuous
or discontinuous mode.
Peak Inductor Current and Fault Current
To ensure that the inductor will not saturate, the peak
inductor current should be calculated knowing the
maximum load current. An appropriate inductor should
then be chosen. In addition, a decision should be made
whether or not the inductor must withstand continuous
fault conditions.
APPLICATIONS INFORMATION
Table 2
VENDOR/
PART NO.
VALUE
(μH)
I
DC
(AMPS)
DCR
(OHMS)
HEIGHT
(mm)
Coiltronics
CTX15-1P 15 1.4 0.087 4.2
CTX15-1 15 1.1 0.08 4.2
CTX33-2P 33 1.3 0.126 6
CTX33-2 33 1.4 0.106 6
UP2-330 33 2.4 0.099 5.9
UP2-470 47 1.9 0.146 5.9
UP2-680 68 1.7 0.19 5.9
UP2-101 100 1.4 0.277 5.9
Sumida
CDRH6D28-150M 15 1.4 0.076 3
CDRH6D38-150M 15 1.6 0.062 4
CDRH6D28-330M 33 0.97 0.122 3
CDRH104R-330M 33 2.1 0.069 3.8
CDRH125-330M 33 2.1 0.044 6
CDRH104R-470M 47 2.1 0.095 3.8
CDRH125-470M 47 1.8 0.058 6
CDRH6D38-680M 68 0.75 0.173 4
CDRH104R-680M 68 1.5 0.158 3.8
CDRH125-680M 68 1.5 0.093 6
CDRH104R-101M 100 1.35 0.225 3.8
CDRH125-101M 100 1.3 0.120 6
Coilcraft
DT3316P-153 15 1.8 0.06 5
DT3316P-333 33 1.3 0.09 5
DT3316P-473 47 1 0.11 5
P1-P3P4-P6P7-P9P10-P12P13-P15P16-P18P19-P21P22-P24P25-P27P28-P30

LT1766EGN-5#PBF

Mfr. #:
Buy LT1766EGN-5#PBF
Manufacturer:
Analog Devices / Linear Technology
Description:
Switching Voltage Regulators 1.5A 200kHz High Voltage Step-down Regulator
Lifecycle:
New from this manufacturer.
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