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LT3759HMSE#PBF

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LT3759
13
3759fc
For more information www.linear.com/3759
Due to the current limit function of the SENSE pin, R
SENSE
should be selected to guarantee that the peak current sense
voltage V
SENSE(PEAK)
during steady state normal opera-
tion is lower than the SENSE current limit threshold (see
the Electrical Characteristics table). Given a 20% margin,
V
SENSE(PEAK)
is set to be 40mV. Then, the maximum switch
ripple current percentage can be calculated using the fol-
lowing equation:
c =
D
V
SENSE
40mV - 0.5 • DV
SENSE
χ
is used in subsequent design examples to calculate
inductor value. ΔV
SENSE
is the ripple voltage across R
SENSE
.
The LT3759 has internal slope compensation to stabilize
the control loop against sub-harmonic oscillation. When
the LT3759 operates at a high duty cycle in continuous
conduction mode, the SENSE voltage ripple ΔV
SENSE
(re-
fer to Figure 2) needs to be limited to ensure the internal
slope compensation is sufficient to stabilize the control
loop. Figure 3 shows the maximum allowed ΔV
SENSE
over
the duty cycle. It is recommended to check and ensure
ΔV
SENSE
is below the curve at the highest duty cycle.
Figure 4. The RC Filter on SENSE pin
APPLICATIONS INFORMATION
The LT3759 switching controller incorporates 100ns timing
interval to blank the ringing on the current sense signal
immediately after M1 is turned on. This ringing is caused
by the parasitic inductance and capacitance of the PCB
trace, the sense resistor, the diode, and the MOSFET. The
100ns timing interval is adequate for most of the LT3759
applications. In the applications that have very large and
long ringing on the current sense signal, a small RC filter
can be added to filter out the excess ringing. Figure 4
shows the RC filter on SENSE pin. It is usually sufficient
to choose 22Ω for R
FLT
and 2.2nF to 10nF for C
FLT
. Keep
R
FLT’s
resistance low. Remember that there is 50µA (typi-
cal) flowing out of the SENSE pin. Adding R
FLT
will affect
the SENSE current limit threshold:
SENSE _ILIM
FLT
C
FLT
3759 F04
LT3759
R
FLT
R
SENSE
M
1
SENSE
GATE
GND
Figure 3. The Maximum Allowed SENSE Voltage Ripple vs
Duty Cycle
DUTY CYCLE
0
MAXIMUM ∆V
SENSE
(mV)
40
30
60
50
1
3759 F03
20
0.1
0.5 0.6 0.7
0.2 0.3 0.4
0.8 0.9
10
0
APPLICATION CIRCUITS
The LT3759 can be configured as different topologies.
The design procedure for component selection differs
somewhat between these topologies. The first topology
to be analyzed will be the boost converter, followed by the
flyback SEPIC and inverting converters.
Boost Converter: Switch Duty Cycle and Frequency
The LT3759 can be configured as a boost converter for
the applications where the converter output voltage is
higher than the input voltage. Remember that boost con-
verters are not short-circuit protected. Under a shorted
output condition, the inductor current is limited only by
the input supply capability. For applications requiring a
step-up converter that is short-circuit protected, please
refer to the Applications Information section covering
SEPIC converters.
LT3759
14
3759fc
For more information www.linear.com/3759
The selection of switching frequency is the starting point.
The maximum frequency that can be used is based on the
maximum duty cycle. The conversion ratio as a function
of duty cycle is:
V
OUT
V
IN
=
1
1−D
in continuous conduction mode (CCM). The equations
that follow assume CCM operation.
For a boost converter operating in CCM, the duty cycle
of the main switch can be calculated based on the output
voltage (V
OUT
) and the input voltage (V
IN
). The maximum
duty cycle (D
MAX
) occurs when the converter has the
minimum input voltage:
D
MAX
=
V
OUT
− V
IN(MIN)
V
OUT
The alternative to CCM, discontinuous conduction mode
(DCM) is not limited by duty cycle to provide high con-
version ratios at a given frequency. The price one pays
is reduced efficiency and substantially higher switching
current.
Boost Converter: Inductor and Sense Resistor Selection
For the boost topology, the maximum average inductor
current is:
I
L(MAX)
= I
O(MAX)
•
1
1−D
MAX
Then, the ripple current can be calculated by:
DI
L
= c • I
L(MAX)
= c • I
O(MAX)
•
1
1- D
MAX
The constant
χ
in the preceding equation represents the
percentage peak-to-peak ripple current in the inductor,
relative to I
L(MAX)
.
The inductor ripple current has a direct effect on the choice
of inductor value. Choosing smaller values of ΔI
L
requires
large inductances and reduces the current loop gain (the
converter will approach voltage mode). Accepting larger
values of ΔI
L
provides fast transient response and allows
the use of low inductances, but results in higher input
current ripple and greater core losses. It is recommended
that
χ
falls within the range of 0.2 to 0.6.
The peak and RMS inductor current are:
I
L(PEAK)
= I
L(MAX)
• 1+
χ
2
I
L(RMS)
= I
L(MAX)
• 1+
χ
2
12
The inductor used with the LT3759 should have a saturation
current rating appropriate to the maximum switch current
selected with the R
SENSE
resistor. Choose an inductor value
based on operating frequency, input and output voltage
to provide a current mode ramp on SENSE during the
switch on-time of approximately 10mV magnitude. The
following equation is useful to estimate the inductor value
for continuous conduction mode operation:
L =
R
SENSE
• V
IN(MIN)
0.01V • f
OSC
• D
MAX
Set the sense voltage at I
L(PEAK)
to be the minimum of the
SENSE current limit threshold with a 20% margin. The
sense resistor value can then be calculated to be:
R
SENSE
=
40mV
I
L(PEAK)
Boost Converter: Power MOSFET Selection
Important parameters for the power MOSFET include the
drain-source voltage rating (V
DS
), the threshold voltage
(V
GS(TH)
), the on-resistance (R
DS(ON)
), the gate to source
and gate to drain charges (Q
GS
and Q
GD
), the maximum
drain current (I
D(MAX)
) and the MOSFET’s thermal resis-
tances (R
θJC
and R
θJA
).
The power MOSFET will see full output voltage, plus a
diode forward voltage, and any additional ringing across
its drain-to-source during its off-time. It is recommended
APPLICATIONS INFORMATION
LT3759
15
3759fc
For more information www.linear.com/3759
to choose a MOSFET whose BV
DSS
is higher than V
OUT
by a safety margin (a 10V safety margin is usually
sufficient).
The power dissipated by the MOSFET in a boost
converter is:
P
FET
= I
2
L(MAX)
•R
DS(ON)
•
D
MAX
+V
2
OUT
•
I
L(MAX)
•
C
RSS
•
f
1A
The first term in the preceding equation represents the
conduction losses in the devices, and the second term, the
switching loss. C
RSS
is the reverse transfer capacitance,
which is usually specified in the MOSFET characteristics.
For maximum efficiency, R
DS(ON)
and C
RSS
should be
minimized. From a known power dissipated in the power
MOSFET, its junction temperature can be obtained using
the following equation:
T
J
=
T
A
•
P
F ET
•
θ
JA
= T
A
+P
FET
•(θ
J C
+θ
CA
)
T
J
must not exceed the MOSFET maximum junction
temperature rating. It is recommended to measure the
MOSFET temperature in steady state to ensure that absolute
maximum ratings are not exceeded.
Boost Converter: Output Diode Selection
To maximize efficiency, a fast switching diode with low
forward drop and low reverse leakage is desirable. The
peak reverse voltage that the diode must withstand is
equal to the regulator output voltage plus any additional
ringing across its anode-to-cathode during the on-time.
The average forward current in normal operation is equal
to the output current, and the peak current is equal to:
I
D(PEAK)
= I
L(PEAK)
= 1+
χ
2
•I
L(MAX)
It is recommended that the peak repetitive reverse voltage
rating V
RRM
is higher than V
OUT
by a safety margin (a 10V
safety margin is usually sufficient).
The power dissipated by the diode is:
P
D
= I
O(MAX)
• V
D
and the diode junction temperature is:
T
J
=
T
A
•
P
D
•
R
θJ A
The R
θJA
to be used in this equation normally includes the
R
θJC
for the device plus the thermal resistance from the
board to the ambient temperature in the enclosure. T
J
must
not exceed the diode maximum junction temperature rating.
Boost Converter: Output Capacitor Selection
Contributions of ESR (equivalent series resistance), ESL
(equivalent series inductance) and the bulk capacitance
must be considered when choosing the correct output
capacitors for a given output ripple voltage. The effect of
these three parameters (ESR, ESL and bulk C) on the output
voltage ripple waveform for a typical boost converter is
illustrated in Figure 5.
The choice of component(s) begins with the maximum
APPLICATIONS INFORMATION
Figure 5. The Output Ripple Waveform of a Boost Converter
V
OUT
(AC)
t
ON
∆V
ESR
RINGING DUE TO
TOTAL INDUCTANCE
(BOARD + CAP)
∆V
COUT
3759 F05
t
OFF
acceptable ripple voltage (expressed as a percentage of
the output voltage), and how this ripple should be divided
between the ESR step ΔV
ESR
and charging/discharging
ΔV
COUT
. For the purpose of simplicity, we will choose
2% for the maximum output ripple, to be divided equally
between ΔV
ESR
and ΔV
COUT
. This percentage ripple will
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LT3759HMSE#PBF

Mfr. #:
Buy LT3759HMSE#PBF
Manufacturer:
Analog Devices / Linear Technology
Description:
Switching Voltage Regulators Boost, Flyback, SEPIC, and Inverting Controller
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