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LTC3727AIG-1#TRPBF

P1-P3P4-P6P7-P9P10-P12P13-P15P16-P18P19-P21P22-P24P25-P27P28-P30P31-P32
LTC3727A-1
16
3727a1fa
The benefi t of the LTC3727A-1 multiphase can be calculated
by using the equation above for the higher power control-
ler and then calculating the loss that would have resulted
if both controller channels switch on at the same time.
The total RMS power lost is lower when both controllers
are operating due to the interleaving of current pulses
through the input capacitor’s ESR. This is why the input
capacitor’s requirement calculated above for the worst-
case controller is adequate for the dual controller design.
Remember that input protection fuse resistance, battery
resistance and PC board trace resistance losses are also
reduced due to the reduced peak currents in a multiphase
system. The overall benefi t of a multiphase design will only
be fully realized when the source impedance of the power
supply/battery is included in the effi ciency testing. The
drains of the two top MOSFETS should be placed within
1cm of each other and share a common C
IN
(s). Separating
the drains and C
IN
may produce undesirable voltage and
current resonances at V
IN
.
The selection of C
OUT
is driven by the required effective
series resistance (ESR). Typically once the ESR require-
ment is satisfi ed the capacitance is adequate for fi ltering.
The output ripple (ΔV
OUT
) is determined by:
ΔΔV I ESR
fC
OUT L
OUT
≅+
⎛
⎝
⎜
⎞
⎠
⎟
1
8
Where f = operating frequency, C
OUT
= output capacitance,
and ΔI
L
= ripple current in the inductor. The output ripple
is highest at maximum input voltage since ΔI
L
increases
with input voltage. With ΔI
L
= 0.3I
OUT(MAX)
the output ripple
will typically be less than 50mV at max V
IN
assuming:
C
OUT
Recommended ESR < 2 R
SENSE
and C
OUT
> 1/(8fR
SENSE
)
The fi rst condition relates to the ripple current into the ESR
of the output capacitance while the second term guarantees
that the output capacitance does not signifi cantly discharge
during the operating frequency period due to ripple current.
The choice of using smaller output capacitance increases
APPLICATIONS INFORMATION
the ripple voltage due to the discharging term but can be
compensated for by using capacitors of very low ESR to
maintain the ripple voltage at or below 50mV. The I
TH
pin
OPTI-LOOP compensation components can be optimized
to provide stable, high performance transient response
regardless of the output capacitors selected.
Manufacturers such as Nichicon, Nippon Chemi-Con and
Sanyo can be considered for high performance through-
hole capacitors. The OS-CON semiconductor dielectric
capacitor available from Sanyo has the lowest (ESR)(size)
product of any aluminum electrolytic at a somewhat
higher price. An additional ceramic capacitor in parallel
with OS-CON capacitors is recommended to reduce the
inductance effects.
In surface mount applications multiple capacitors may
need to be used in parallel to meet the ESR, RMS current
handling and load step requirements of the application.
Aluminum electrolytic, dry tantalum and special polymer
capacitors are available in surface mount packages. Special
polymer surface mount capacitors offer very low ESR but
have lower storage capacity per unit volume than other
capacitor types. These capacitors offer a very cost-effective
output capacitor solution and are an ideal choice when
combined with a controller having high loop bandwidth.
Tantalum capacitors offer the highest capacitance density
and are often used as output capacitors for switching
regulators having controlled soft-start. Several excellent
surge-tested choices are the AVX TPS, AVX TPS Series
III or the KEMET T510 series of surface mount tantalums,
available in case heights ranging from 1.2mm to 4.1mm.
Aluminum electrolytic capacitors can be used in cost-
driven applications providing that consideration is given
to ripple current ratings, temperature and long term
reliability. A typical application will require several to many
aluminum electrolytic capacitors in parallel. A combination
of the above mentioned capacitors will often result in
maximizing performance and minimizing overall cost. Other
capacitor types include Nichicon PL series, NEC Neocap,
Cornell Dubilier ESRE and Sprague 595D series. Consult
manufacturers for other specifi c recommendations.
LTC3727A-1
17
3727a1fa
INTV
CC
Regulator
An internal P-channel low dropout regulator produces 7.5V
at the INTV
CC
pin from the V
IN
supply pin. INTV
CC
powers
the drivers and internal circuitry within the LTC3727A-1.
The INTV
CC
pin regulator can supply a peak current of
50mA and must be bypassed to ground with a minimum
of 4.7μF tantalum, 10μF special polymer, or low ESR type
electrolytic capacitor. A 1μF ceramic capacitor placed di-
rectly adjacent to the INTV
CC
and PGND IC pins is highly
recommended. Good bypassing is necessary to supply
the high transient currents required by the MOSFET gate
drivers and to prevent interaction between channels.
Higher input voltage applications in which large MOSFETs
are being driven at high frequencies may cause the
maximum junction temperature rating for the LTC3727A-1
to be exceeded. The system supply current is normally
dominated by the gate charge current. Additional external
loading of the INTV
CC
and 3.3V linear regulators also
needs to be taken into account for the power dissipation
calculations. The total INTV
CC
current can be supplied by
either the 7.5V internal linear regulator or by the EXTV
CC
input pin. When the voltage applied to the EXTV
CC
pin is
less than 7.3V, all of the INTV
CC
current is supplied by
the internal 7.5V linear regulator. Power dissipation for
the IC in this case is highest: (V
IN
)(I
INTVCC
), and overall
effi ciency is lowered. The gate charge current is dependent
on operating frequency as discussed in the Effi ciency
Considerations section. The junction temperature can be
estimated by using the equations given in Note 2 of the
Electrical Characteristics. For example, the LTC3727A-1
V
IN
current is limited to less than 24mA from a 24V supply
when not using the EXTV
CC
pin as follows:
T
J
= 70°C + (24mA)(24V)(95°C/W) = 125°C
Use of the EXTV
CC
input pin reduces the junction tem-
perature to:
T
J
= 70°C + (24mA)(7.5V)(95°C/W) = 87°C
APPLICATIONS INFORMATION
Dissipation should be calculated to also include any added
current drawn from the internal 3.3V linear regulator.
To prevent maximum junction temperature from being
exceeded, the input supply current must be checked
operating in continuous mode at maximum V
IN
.
EXTV
CC
Connection
The LTC3727A-1 contains an internal P-channel MOSFET
switch connected between the EXTV
CC
and INTV
CC
pins.
When the voltage applied to EXTV
CC
rises above 7.3V,
the internal regulator is turned off and the switch closes,
connecting the EXTV
CC
pin to the INTV
CC
pin thereby sup-
plying internal power. The switch remains closed as long
as the voltage applied to EXTV
CC
remains above 7.0V. This
allows the MOSFET driver and control power to be derived
from the output during normal operation (7.2V < V
OUT
<
8.5V) and from the internal regulator when the output is
out of regulation (start-up, short-circuit). If more current
is required through the EXTV
CC
switch than is specifi ed,
an external Schottky diode can be added between the
EXTV
CC
and INTV
CC
pins. Do not apply greater than 8.5V
to the EXTV
CC
pin and ensure that EXTV
CC
< V
IN
.
Signifi cant effi ciency gains can be realized by powering
INTV
CC
from the output, since the V
IN
current resulting
from the driver and control currents will be scaled by a
factor of (Duty Cycle)/(Effi ciency). For 7.5V regulators
this supply means connecting the EXTV
CC
pin directly to
V
OUT
. However, for 3.3V and other lower voltage regula-
tors, additional circuitry is required to derive INTV
CC
power
from the output.
The following list summarizes the four possible connec-
tions for EXTV
CC
:
1. EXTV
CC
Left Open (or Grounded). This will cause
INTV
CC
to be powered from the internal 7.5V regulator
resulting in an effi ciency penalty of up to 10% at high
input voltages.
LTC3727A-1
18
3727a1fa
2. EXTV
CC
Connected directly to V
OUT
. This is the normal
connection for a 7.5V regulator and provides the highest
effi ciency.
3. EXTV
CC
Connected to an External supply. If an external
supply is available in the 7.5V to 8.5V range, it may be
used to power EXTV
CC
providing it is compatible with the
MOSFET gate drive requirements.
4. EXTV
CC
Connected to an Output-Derived Boost Network.
For 3.3V and other low voltage regulators, effi ciency
gains can still be realized by connecting EXTV
CC
to an
output-derived voltage that has been boosted to greater
than 7.5V. This can be done with the inductive boost
winding as shown in Figure 6.
Topside MOSFET Driver Supply (C
B
, D
B
)
External bootstrap capacitors C
B
connected to the BOOST
pins supply the gate drive voltages for the topside MOSFETs.
Capacitor C
B
in the functional diagram is charged though
external diode D
B
from INTV
CC
when the SW pin is low.
When one of the topside MOSFETs is to be turned on,
the driver places the C
B
voltage across the gate-source
of the desired MOSFET. This enhances the MOSFET and
turns on the topside switch. The switch node voltage, SW,
rises to V
IN
and the BOOST pin follows. With the topside
MOSFET on, the boost voltage is above the input supply:
V
BOOST
= V
IN
+ V
INTVCC
. The value of the boost capacitor
C
B
needs to be 100 times that of the total input capacitance
APPLICATIONS INFORMATION
Figure 6. Secondary Output Loop & EXTV
CC
Connection
EXTV
CC
FCB
SGND
V
IN
TG1
SW
BG1
PGND
LTC3727A-1
R
SENSE
V
OUT
V
SEC
+
C
OUT
+
1μF
3727 F06
N-CH
N-CH
R6
+
C
IN
V
IN
T1
1:N
OPTIONAL EXTV
CC
CONNECTION
7.5V < V
SEC
< 8.5V
R5
of the topside MOSFET(s). The reverse breakdown of the
external Schottky diode must be greater than V
IN(MAX)
.
When adjusting the gate drive level, the fi nal arbiter is the
total input current for the regulator. If a change is made
and the input current decreases, then the effi ciency has
improved. If there is no change in input current, then there
is no change in effi ciency.
Output Voltage
The LTC3727A-1 output voltages are each set by an external
feedback resistive divider carefully placed across the output
capacitor. The resultant feedback signal is compared with
the internal precision 0.800V voltage reference by the error
amplifi er. The output voltage is given by the equation:
VV
R
R
OUT
=+
⎛
⎝
⎜
⎞
⎠
⎟
08 1
2
1
.
where R1 and R2 are defi ned in Figure 2.
SENSE
+
/SENSE
–
Pins
The common mode input range of the current comparator
sense pins is from 0V to 14V. Continuous linear operation is
guaranteed throughout this range allowing output voltage
setting from 0.8V to 14V. A differential NPN input stage is
biased with internal resistors from an internal 2.4V source
as shown in the Functional Diagram. This requires that
current either be sourced or sunk from the SENSE pins
depending on the output voltage. If the output voltage is
below 2.4V current will fl ow out of both SENSE pins to
the main output. The output can be easily preloaded by
the V
OUT
resistive divider to compensate for the current
comparator’s negative input bias current. The maximum
current fl owing out of each pair of SENSE pins is:
I
SENSE
+
+ I
SENSE
–
= (2.4V – V
OUT
)/24k
Since V
OSENSE
is servoed to the 0.8V reference voltage,
we can choose R1 in Figure 2 to have a maximum value
to absorb this current.
Rk
V
VV
VV
MAX
OUT
OUT
124
08
24
24
()
.
.–
.
=
⎛
⎝
⎜
⎞
⎠
⎟
<for
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LTC3727AIG-1#TRPBF

Mfr. #:
Buy LTC3727AIG-1#TRPBF
Manufacturer:
Analog Devices / Linear Technology
Description:
Switching Voltage Regulators Dual, 2-Phase Synchronous Controller w/ up to 14V Output
Lifecycle:
New from this manufacturer.
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