AD600/AD602
Rev. F | Page 11 of 32
It is apparent from the foregoing that it is essential to use a low
resistance in the design of the ladder network to achieve low
noise. In some applications, this can be inconvenient, requiring
the use of an external buffer or preamplifier. However, very few
amplifiers combine the needed low noise with low distortion at
maximum input levels, and the power consumption required to
achieve this performance is quite high (due to the need to
maintain very low resistance values while also coping with large
inputs). On the other hand, there is little value in providing a
buffer with high input impedance because the usual reason for
this—the minimization of loading of a high resistance source—
is not compatible with low noise.
Apart from the small variations just mentioned, the SNR at the
output is essentially independent of the attenuator setting,
because the maximum undistorted output is 1 V rms, and the
NSD at the output of the AD600 is fixed at 113 × 114 nV/√Hz,
or 158 nV/√Hz. Therefore, in a 1 MHz bandwidth, the output
SNR is 76 dB. The input NSD of the AD600/AD602 is the same
but, because of the 10 dB lower gain in the AD602’s fixed
amplifier, its output SNR is 10 dB better, or 86 dB in a 1 MHz
bandwidth.
GAIN-CONTROL INTERFACE
The attenuation is controlled through a differential, high
impedance (15 MΩ) input, with a scaling factor that is laser
trimmed to 32 dB per volt, that is, 31.25 mV/dB. Each of the
two amplifiers has its own control interface. An internal band
gap reference ensures stability of the scaling with respect to
supply and temperature variations and is the only circuitry
common to both channels.
When the differential input voltage V
G
= 0 V, the attenuator
slider is centered, providing an attenuation of +21.07 dB,
resulting in an overall gain of +20 dB (= –21.07 dB + +41.07 dB).
When the control input is −625 mV, the gain is lowered by
+20 dB (= +0.625 × +32) to 0 dB; when set to +625 mV, the
gain is increased by +20 dB to +40 dB. When this interface is
overdriven in either direction, the gain approaches either
−1.07 dB (= −42.14 dB + +41.07 dB) or +41.07 dB (= 0 +
+41.07 dB), respectively.
The gain of the AD600 can be calculated by
Gain (dB) = 32 V
G
+ 20 (1)
where V
G
is in volts.
For the AD602, the expression is
Gain (dB) = 32 V
G
+ 10 (2)
Operation is specified for V
G
in the range from −625 mV dc to
+625 mV dc. The high impedance gain-control input ensures
minimal loading when driving many amplifiers in multiple-
channel applications. The differential input configuration
provides flexibility in choosing the appropriate signal levels
and polarities for various control schemes.
For example, the gain-control input can be fed differentially to
the inputs or single-ended by simply grounding the unused
input. In another example, if the gain is controlled by a DAC
providing a positive-only, ground-referenced output, the gain
control LO pin (either C1LO or C2LO) should be biased to a
fixed offset of 625 mV to set the gain to 0 dB when gain control
HI (C1HI or C2HI) is at zero and to set the gain to 40 dB when
at 1.25 V.
It is a simple matter to include a voltage divider to achieve other
scaling factors. When using an 8-bit DAC with an FS output of
2.55 V (10 mV/bit), a 1.6 divider ratio (generating 6.25 mV/bit)
results in a gain setting resolution of 0.2 dB/bit. The process of
cascading the two sections of an AD600 or AD602 when various
options exist for gain control is explained in the Achieving 80 DB
Gain Range section.
SIGNAL-GATING INPUTS
Each amplifier section of the AD600/AD602 is equipped with a
signal-gating function, controlled by a TTL or CMOS logic
input (GAT1 or GAT2). The ground references for these inputs
are the signal input grounds A1LO and A2LO, respectively.
Operation of the channel is unaffected when this input is LO or
left open-circuited. Signal transmission is blocked when this
input is HI. The dc output level of the channel is set to within a
few millivolts of the output ground (A1CM or A2CM), and
simultaneously the noise level drops significantly. The reduction
in noise and spurious signal feedthrough is useful in ultrasound
beam-forming applications, where many amplifier outputs are
summed.
COMMON-MODE REJECTION
A special circuit technique provides rejection of voltages
appearing between input grounds (A1LO and A2LO) and
output grounds (A1CM and A2CM). This is necessary because
of the op amp form of the amplifier, as shown in Figure 21.
The feedback voltage is developed across the RF1 resistor
(which, to achieve low noise, has a value of only 20 ). The
voltage developed across this resistor is referenced to the input
common, so the output voltage is also referred to that node.
For zero differential signal input between A1HI and A1LO, the
output A1OP simply follows the voltage at A1CM. Note that the
range of voltage differences that can exist between A1LO and
A1CM (or A2LO and A2CM) is limited to about ±100 mV.
Figure 18 shows the typical common-mode rejection ratio vs.
frequency.
ACHIEVING 80 dB GAIN RANGE
The two amplifier sections of the X-AMP can be connected in
series to achieve higher gain. In this mode, the output of A1
(A1OP and A1CM) drives the input of A2 via a high-pass
network (usually just a capacitor) that rejects the dc offset.
The nominal gain range is now –2 dB to +82 dB for the AD600
or −22 dB to +62 dB for the AD602.
