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LT3844IFE#PBF

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LT3844
19
3844fc
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applicaTions inForMaTion
Minimum On-Time Considerations (Buck Mode)
Minimum on-time, t
ON(MIN)
, is the smallest amount of time
that the LT3844 is capable of turning the top MOSFET on
and off again. It is determined by internal timing delays
and the amount of gate charge required turning on the
top MOSFET. Low duty cycle applications may approach
this minimum on-time limit and care should be taken to
ensure that:
t
ON
=
V
OUT
V
IN
• f
SW
> t
ON(MIN)
where t
ON(MIN)
is typically 350ns worst case.
If the duty cycle falls below what can be accommodated
by the minimum on-time, the LT3844 will begin to skip
cycles. The output will be regulated, but the ripple current
and ripple voltage will increase. If lower frequency opera
-
tion is acceptable, the on-time can be increased above
t
ON(MIN)
for the same step-down ratio.
Boost Converter Design
The LT3844 can be used to configure a boost converter
to step-up voltages to as high as hundreds of volts. An
example of a boost converter circuit schematic is shown
in the Typical Applications section. The following sections
are a guide to designing a boost converter:
The maximum duty cycle of the main switch is:
DC
MAX
=
V
OUT
−
V
IN(MIN)
V
OUT
Boost Converter: Inductor Selection
The critical parameters for selection of an inductor are
minimum inductance value and saturation current. The
minimum inductance value is calculated as follows:
L
MIN
=
V
IN(MIN)
∆I
L
• f
SW
• DC
MAX
f
SW
is the switch frequency.
Similar to the buck converter, the typical range of values
for ∆I
L
is (0.2 • I
L(MAX)
) to (0.5 • I
L(MAX)
), where I
L(MAX)
is the maximum average inductor current.
I
L(MAX)
=I
OUT(MAX)
•
V
OUT
V
IN(MIN)
Using ∆I
L
= 0.3 • I
L(MAX)
yields a good design compromise
between inductor performance versus inductor size and
cost.
The inductor must not saturate at the peak operating
current, I
L(MAX)
+ ∆I
L
/2. The inductor saturation current
specification is the current at which the inductance, mea-
sured at zero current, decreases by a specified amount,
typically 30%.
One drawback of boost regulators is that they cannot be
current limited for output shorts because the current steer
-
ing diode makes a direct connection between input and
output. Therefore, the inductor current during an output
short
circuit is only limited by the available current of the
input supply.
After calculating the minimum inductance value and the
saturation current for your design, select an off-the-shelf
inductor. For more detailed information on selecting an
inductor, please see the “Inductor Selection” section of
Linear Technology Application Note 19.
Boost Converter: MOSFET Selection
The selection criteria of the external N-channel standard
level power MOSFET include on resistance (R
DS(ON)
), re-
verse transfer capacitance (C
RSS
), maximum drain source
voltage (V
DSS
), total gate charge (Q
G
) and maximum
continuous drain current.
For maximum efficiency, minimize R
DS(ON)
and C
RSS
.
Low R
DS(ON)
minimizes conduction losses while low C
RSS
minimizes transition losses. The problem is that R
DS(ON)
is
inversely related to C
RSS
. Balancing the transition losses
with the conduction losses is a good idea in sizing the
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MOSFET. Select the MOSFET to balance the two losses.
Calculate the maximum conduction losses of the MOSFET:
P
COND
= DC
MAX
I
OUT(MAX)
1−DC
MAX
⎛
⎝
⎜
⎞
⎠
⎟
• R
DS(ON)
Note that R
DS(ON)
has large positive temperature depen-
dence. The MOSFET manufacturer’s data sheet contains
a curve, R
DS(ON)
vs Temperature. Calculate the maximum
transition losses:
P
TRAN
=
(k)(V
OUT
)
2
(I
OUT(MAX)
)(C
RSS
)(f
SW
)
(1−DC
MAX
)
where k is a constant inversely related to the gate driver
current, approximated by k = 2 for LT3844 applications.
The total maximum power dissipation of the MOSFET is
the sum of these two loss terms:
P
FET(TOTAL)
= P
COND
+ P
TRAN
To achieve high supply efficiency, keep the P
FET(TOTAL)
to
less than 3% of the total output power. Also, complete
a thermal analysis to ensure that the MOSFET junction
temperature is not exceeded.
T
J
= T
A
+ P
FET(TOTAL)
• θ
JA
where θ
JA
is the package thermal resistance and T
A
is the
ambient temperature. Keep the calculated T
J
below the
maximum specified junction temperature, typically 150°C.
Note that when V
OUT
is high (>20V), the transition losses
may dominate. A MOSFET with higher R
DS(ON)
and lower
C
RSS
may provide higher efficiency. MOSFETs with higher
voltage V
DSS
specification usually have higher R
DS(ON)
and lower C
RSS
.
Choose the MOSFET V
DSS
specification to exceed the
maximum voltage across the drain to the source of the
MOSFET, which is V
OUT
plus the forward voltage of the
rectifier, typically less than 1V.
The internal V
CC
regulator is capable of sourcing up to
40mA which limits the maximum total MOSFET gate
charge, Q
G
, to 40mA / f
SW
. The Q
G
vs V
GS
specification
is typically provided in the MOSFET data sheet. Use Q
G
at
V
GS
of 8V. If V
CC
is back driven from an external supply,
the MOSFET drive current is not sourced from the internal
regulator of the LT3844 and the Q
G
of the MOSFET is not
limited by the IC. However, note that the MOSFET drive
current is supplied by the internal regulator when the
external supply back driving V
CC
is not available such as
during start-up or short-circuit.
The manufacturer’s maximum continuous drain current
specification should exceed the peak switch current which
is the same as the inductor peak current, I
L(MAX)
+ ∆I
L
/2.
During the supply start-up, the gate drive levels are set by
the V
CC
voltage regulator, which is approximately 8V. Once
the supply is up and running, the V
CC
can be back driven
by an auxiliary supply such as V
OUT
. It is important not to
exceed the manufacturer’s maximum V
GS
specification.
A standard level threshold MOSFET typically has a V
GS
maximum of 20V.
Boost Converter: Rectifier Selection
The rectifier is selected based upon the forward voltage,
reverse voltage and maximum current. A Schottky diode
is recommended for its low forward voltage and yields the
lowest power loss and highest efficiency. The maximum
reverse voltage that the diode will see is V
OUT
. The average
diode current is equal to the maximum output load current,
I
OUT(MAX)
. A diode rated at 1.5 to 2 times the maximum
average diode current is recommended. Remember boost
converters are not short-circuit protected.
Boost Converter: Output Capacitor Selection
In boost mode, the output capacitor requirements are
more demanding due to the fact that the current waveform
is pulsed instead of continuous as in a buck converter.
The choice of component(s) is driven by the acceptable
ripple voltage which is affected by the ESR, ESL and bulk
capacitance. The total output ripple voltage is:
∆V
OUT
=I
OUT(MAX)
1
f
SW
• C
OUT
+
ESR
1−DC
MAX
⎛
⎝
⎜
⎜
⎞
⎠
⎟
⎟
where the first term is due to the bulk capacitance and the
second term due to the ESR.
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The choice of output capacitor is also driven by the RMS
ripple current requirement. The RMS ripple current is:
I
RMS(COUT)
=I
OUT(MAX)
•
V
OUT
− V
IN(MIN)
V
IN(MIN)
At lower output voltages (<30V) it may be possible to sat-
isfy both the output ripple voltage and RMS requirements
with one or more capacitors of a single type. However, at
output voltages above
30V where capacitors with both low
ESR and high bulk capacitance are hard to find, the best
approach is to use a combination of aluminum electrolytic
and ceramic capacitors. The low ESR ceramic capacitor
will minimize the ESR while the Aluminum Electrolytic
capacitor will supply the required bulk capacitance.
Boost Converter: Input Capacitor Selection
The input capacitor of a boost converter is less critical
than the output capacitor, due to the fact that the inductor
is in series with the input and the input current waveform
is continuous. The input voltage source impedance de
-
termines the size of the input capacitor, which is typically
in the range of 10µF
to 100µF. A low ESR capacitor is
recommended though not as critical as with the output
capacitor. The RMS input capacitor ripple current for a
boost converter is:
I
RMS(CIN)
= 0.3 •
V
IN(MIN)
L • f
SW
• DC
MAX
Please note that the input capacitor can see a very high
surge current when a battery is suddenly connected to
the input of the converter and solid tantalum capacitors
can fail catastrophically under these conditions. Be sure
to specify surge-tested capacitors.
Boost Converter: R
SENSE
Selection
The boost application in the Typical Applications section
has the location of the current sense resistor in series with
the inductor with one side referenced to V
IN
. This location
was chosen for two reasons. Firstly, the circulating current
is always monitored so in the case of an output overvoltage
or input overcurrent condition the main switch will skip
cycles to protect the circuitry. Secondly, the V
IN
node can
be considered low noise since it is heavily filtered and the
input current is not pulsed but continuous.
In the case where the input voltage exceeds the voltage
limits on the LT3844 Sense pins, the sense resistor can
be moved to the source of the MOSFET. In both cases the
resistor value is the calculated using the same formula.
The LT3844 current comparator has a maximum threshold
of 100mV/R
SENSE
. The current comparator threshold sets
the peak of the inductor current. Allowing adequate margin
for ripple current and external component tolerances,
R
SENSE
can be calculated as follows:
R
SENSE
=
70mV
I
L(MAX)
Where I
L(MAX)
is the maximum average inductor current
as calculated in the Boost Converter: Inductor Selection
section.
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LT3844IFE#PBF

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Switching Voltage Regulators 60V DC/DC Controller w/ PLL
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