LTC3832-1ES8 Datasheet LTC3832-1ES8#TRPBF, LTC3832EGN#TRPBF, LTC3832-1ES8#PBF | P13-P15

LTC3832/LTC3832-1

sn3832 3832fs

Diagram). This increases the G2 on-time and allows the

charge pump capacitors to be refreshed.

For applications using an external supply to PV

CC1

, this

supply must also be higher than V

by at least 2.5V to

ensure normal operation.

For applications with a 5V or higher V

supply, PV

CC2

can

be tied to V

if a logic level MOSFET is used. PV

CC1

can be

supplied using a doubling charge pump as shown in

Figure␣ 9. This circuit provides 2V

– V

to PV

CC1

while Q1

is ON.

enhance standard power MOSFETs. Under this condition,

the effective MOSFET R

DS(ON)

may be quite high, raising

the dissipation in the FETs and reducing efficiency. Logic

level FETs are the recommended choice for 5V or lower

voltage systems. Logic level FETs can be fully enhanced

with a doubler/tripling charge pump and will operate at

maximum efficiency.

After the MOSFET threshold voltage is selected, choose the

DS(ON)

based on the input voltage, the output voltage,

allowable power dissipation and maximum output current.

In a typical LTC3832 circuit, operating in continuous mode,

the average inductor current is equal to the output load

current. This current flows through either Q1 or Q2 with the

power dissipation split up according to the duty cycle:

DC Q

OUT

IN OUT

()

() –

–

The R

DS(ON)

required for a given conduction loss can now

be calculated by rearranging the relation P = I

DC Q I

VV I

DS ON Q

MAX Q

LOAD

IN MAX Q

OUT LOAD

DS ON Q

MAX Q

LOAD

IN MAX Q

IN OUT LOAD

()

() ()

()

() ()

()•( )

•

•( )

()•( )

•

(– )•( )

MAX

should be calculated based primarily on required

efficiency or allowable thermal dissipation. A typical high

efficiency circuit designed for 3.3V input and 2.5V at 10A

output might allow no more than 3% efficiency loss at full

load for each MOSFET. Assuming roughly 90% efficiency

at this current level, this gives a P

MAX

value of:

(2.5V)(10A/0.9)(0.03) = 0.83W per FET

and a required R

DS(ON)

of:

VVA

DS ON Q

()

(. )•(. )

( . )( )

(. )•(. )

( . – . )( )

33 083

25 10

0 011

33 083

33 25 10

0 034

==Ω

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LTC3832

3832 F09

12V

1N5242

Q2 C

OUT

0.1µF

CC2

OPTIONAL

USE FOR V

≥ 7V

MBR0530T1

CC1

Figure 9. Doubling Charge Pump

Power MOSFETs

Two N-channel power MOSFETs are required for most

LTC3832 circuits. These should be selected based

primarily on threshold voltage and on-resistance consid-

erations. Thermal dissipation is often a secondary con-

cern in high efficiency designs. The required MOSFET

threshold should be determined based on the available

power supply voltages and/or the complexity of the gate

drive charge pump scheme. In 3.3V input designs where

an auxiliary 12V supply is available to power PV

CC1

and

CC2

, standard MOSFETs with R

DS(ON)

specified at V

= 5V or 6V can be used with good results. The current

drawn from this supply varies with the MOSFETs used

and the LTC3832’s operating frequency, but is generally

less than 50mA.

LTC3832 applications that use 5V or lower V

voltage and

a doubling/tripling charge pump to generate PV

CC1

and

CC2

, do not provide enough gate drive voltage to fully

LTC3832/LTC3832-1

sn3832 3832fs

Note that the required R

DS(ON)

for Q2 is roughly three

times that of Q1 in this example. Note also that while the

required R

DS(ON)

values suggest large MOSFETs, the

power dissipation numbers are only 0.83W per device or

less; large TO-220 packages and heat sinks are not neces-

sarily required in high efficiency applications. Siliconix

Si4410DY or International Rectifier IRF7413 (both in

SO-8) or Siliconix SUD50N03-10 (TO-252) or ON Semi-

conductor MTD20N03HDL (DPAK) are small footprint

surface mount devices with R

DS(ON)

values below 0.03Ω

at 5V of V

that work well in LTC3832 circuits. Using a

higher P

MAX

value in the R

DS(ON)

calculations generally

decreases the MOSFET cost and the circuit efficiency and

increases the MOSFET heat sink requirements.

Table 1 highlights a variety of power MOSFETs for use in

LTC3832 applications.

Inductor Selection

The inductor is often the largest component in an LTC3832

design and must be chosen carefully. Choose the inductor

value and type based on output slew rate requirements. The

maximum rate of rise of inductor current is set by the

inductor’s value, the input-to-output voltage differential and

the LTC3832’s maximum duty cycle. In a typical 3.3V in-

put, 2.5V output application, the maximum rise time will be:

DC V V

MAX IN OUT

•( – ) .

076

where L

is the inductor value in µH. With proper fre-

quency compensation, the combination of the inductor

and output capacitor values determine the transient recov-

ery time. In general, a smaller value inductor improves

transient response at the expense of ripple and inductor

core saturation rating. A 1µH inductor has a 0.76A/µs rise

time in this application, resulting in a 6.6µs delay in

responding to a 5A load current step. During this 6.6µs,

the difference between the inductor current and the output

current is made up by the output capacitor. This action

causes a temporary voltage droop at the output. To

minimize this effect, the inductor value should usually be

in the 1µH to 5µH range for most 3.3V input LTC3832

circuits. To optimize performance, different combinations

of input and output voltages and expected loads may

require different inductor values.

Once the required value is known, the inductor core type

can be chosen based on peak current and efficiency

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Table 1. Recommended MOSFETs for LTC3832 Applications

TYPICAL INPUT

DS(ON)

CAPACITANCE

PARTS AT 25°C (mΩ) RATED CURRENT (A) C

ISS

(pF) θ

(°C/W) T

JMAX

(°C)

Siliconix SUD50N03-10 19 15 at 25°C 3200 1.8 175

TO-252 10 at 100°C

Siliconix Si4410DY 20 10 at 25°C 2700 150

SO-8 8 at 70°C

ON Semiconductor MTD20N03HDL 35 20 at 25°C 880 1.67 150

DPAK 16 at 100°C

Fairchild FDS6670A 8 13 at 25°C 3200 25 150

S0-8

Fairchild FDS6680 10 11.5 at 25°C 2070 25 150

SO-8

ON Semiconductor MTB75N03HDL 9 75 at 25°C 4025 1 150

DD PAK 59 at 100°C

IR IRL3103S 19 64 at 25°C 1600 1.4 175

DD PAK 45 at 100°C

IR IRLZ44 28 50 at 25°C 3300 1 175

TO-220 36 at 100°C

Fuji 2SK1388 37 35 at 25°C 1750 2.08 150

TO-220

Note: Please refer to the manufacturer’s data sheet for testing conditions and detailed information.

LTC3832/LTC3832-1

sn3832 3832fs

requirements. Peak current in the inductor will be equal to

the maximum output load current plus half of the peak-to-

peak inductor ripple current. Ripple current is set by the

inductor value, the input and output voltage and the

operating frequency. The ripple current is approximately

equal to:

VV V

fLV

RIPPLE

IN OUT OUT

OSC O IN

−()•()

••

OSC

= LTC3832 oscillator frequency = 300kHz

= Inductor value

Solving this equation with our typical 3.3V to 2.5V appli-

cation with a 1µH inductor, we get:

(. – . )• .

••.

33 25 25

300 1 3 3

VVV

kHz H V

-P

Peak inductor current at 10A load:

10A + (2A/2) = 11A

The ripple current should generally be between 10% and

40% of the output current. The inductor must be able to

withstand this peak current without saturating, and the

copper resistance in the winding should be kept as low as

possible to minimize resistive power loss. Note that in

circuits not employing the current limit function, the

current in the inductor may rise above this maximum

under short-circuit or fault conditions; the inductor should

be sized accordingly to withstand this additional current.

Inductors with gradual saturation characteristics are often

the best choice.

Input and Output Capacitors

A typical LTC3832 design places significant demands on

both the input and the output capacitors. During normal

steady load operation, a buck converter like the LTC3832

draws square waves of current from the input supply at the

switching frequency. The peak current value is equal to the

output load current plus 1/2 the peak-to-peak ripple cur-

rent. Most of this current is supplied by the input bypass

capacitor. The resulting RMS current flow in the input

capacitor heats it and causes premature capacitor failure

in extreme cases. Maximum RMS current occurs with

50% PWM duty cycle, giving an RMS current value equal

to I

OUT

/2. A low ESR input capacitor with an adequate

ripple current rating must be used to ensure reliable

operation. Note that capacitor manufacturers’ ripple cur-

rent ratings are often based on only 2000 hours (3 months)

lifetime at rated temperature. Further derating of the input

capacitor ripple current beyond the manufacturer’s speci-

fication is recommended to extend the useful life of the

circuit. Lower operating temperature has the largest effect

on capacitor longevity.

The output capacitor in a buck converter under steady-

state conditions sees much less ripple current than the

input capacitor. Peak-to-peak current is equal to inductor

ripple current, usually 10% to 40% of the total load

current. Output capacitor duty places a premium not on

power dissipation but on ESR. During an output load

transient, the output capacitor must supply all of the

additional load current demanded by the load until the

LTC3832 adjusts the inductor current to the new value.

ESR in the output capacitor results in a step in the output

voltage equal to the ESR value multiplied by the change in

load current. An 5A load step with a 0.05Ω ESR output

capacitor results in a 250mV output voltage shift; this is

10% of the output voltage for a 2.5V supply! Because of

the strong relationship between output capacitor ESR and

output load transient response, choose the output capaci-

tor for ESR, not for capacitance value. A capacitor with

suitable ESR will usually have a larger capacitance value

than is needed to control steady-state output ripple.

Electrolytic capacitors rated for use in switching power

supplies with specified ripple current ratings and ESR can

be used effectively in LTC3832 applications. OS-CON

electrolytic capacitors from Sanyo and other manufactur-

ers give excellent performance and have a very high

performance/size ratio for electrolytic capacitors. Surface

mount applications can use either electrolytic or dry

tantalum capacitors. Tantalum capacitors must be surge

tested and specified for use in switching power supplies.

Low cost, generic tantalums are known to have very short

lives followed by explosive deaths in switching power

supply applications. Other capacitors that can be used

include the Sanyo POSCAP and MV-WX series.

A common way to lower ESR and raise ripple current

capability is to parallel several capacitors. A typical

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Description:

Switching Voltage Regulators LTC3832 - High Power Step-Down Synchronous DC/DC Controllers for Low Voltage Operation

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LTC3832-1ES8

LTC3832-1ES8

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