A6265KLPTR-T Datasheet A6265KLPTR-T | P7-P9

Automotive High Current LED Controller

A6265

Allegro MicroSystems, LLC

115 Northeast Cutoff

Worcester, Massachusetts 01615-0036 U.S.A.

1.508.853.5000; www.allegromicro.com

Functional Description

The A6265 is a DC-DC converter controller that is designed to

drive series-connected high power LEDs in automotive applica-

tions. It provides programmable constant current output at load

voltages and currents limited only by the external components.

For automotive applications optimum performance is achieved

when driving between two and ten LEDs at currents up to 1 A.

The A6265 can be configured as a standard boost converter or

as a supply referenced boost converter. In the supply referenced

configuration the load voltage is the difference between the boost

voltage and the supply voltage. This difference can be greater

than, equal to, or less than the supply voltage, effectively provid-

ing a buck-boost capability. This configuration provides seam-

less, uninterrupted operation over the wide supply voltage range

possible in automotive applications and, because the output is ref-

erenced to the positive supply, there is no load current to ground.

This ensures that there is no leakage path through the LEDs when

in shutdown and no inrush current at power-up.

The A6265 integrates all necessary control elements to pro-

vide a cost-effective solution using a single external logic-level

MOSFET and minimum additional external passive components.

The LED current is set by selecting an appropriate value for the

sense resistor value and using the EN input to provide simple

on-off control or for PWM brightness control using a suitable

externally generated PWM signal. The LED current can be

reduced in a single step by reducing the voltage between the

IREF pin and GND to less than 1 V.

The pin functions and circuit operation are described in detail in

the following sections.

Pin Functions

VIN Supply to the control circuit. A bypass capacitor must be

connected between this pin and GND.

GND Ground reference connection. This pin should be connected

directly to the negative supply.

EN Logic input to enable operation. Can be used as direct PWM

input. Chip enters low power sleep mode when low for longer

than the disable time, t

DIS

FF1 Fault Flag output and isolation control. Open drain current

sink output, when high impedance indicates detection of a critical

circuit fault. An external pull-up resistor should be connected to a

suitable logic supply for simple logic fault flag operation or to the

source of the PMOS FET used to isolate the load from the supply.

Table 1 defines when FF1 is active. If FF1 is pulled low when

an output short fault is indicated then the output disable will be

overridden.

FF2 Fault Flag output. Open drain output, when high impedance

indicates detection of a circuit fault. An external pull-up resistor

should be connected to a suitable logic supply. If VREG is not

used, then the logic supply should not be pulled 300 mV above

VREG. Table 1 defines when FF2 is active. If FF2 is pulled low

when an open LED fault is indicated then the output disable will

be overridden.

OSC Resistor to ground to set the internal oscillator or clock

input from external oscillator. When connected to VREG the

oscillator runs at typically 350 kHz. Higher accuracy in the

frequency is possible by connecting a resistor from this pin to

ground or by driving this pin with an external precision oscillator.

CKOUT Logic output at the oscillator frequency with phase

shift. Used to drive succeeding controllers to interleave switching

instants.

IREF LED current reference modifier. A voltage input that can be

used to reduce the LED current sense voltage. When connected to

VREG, the current sense voltage, V

IDL

, and the value of the sense

resistor, R

, define the maximum LED current.

SG Gate drive for external logic-level MOSFET low-side switch

that connects the inductor to ground.

SP, SN Sense amplifier connections for switch current limit

sense resistor, R

LP Positive sense amplifier connection for LED current limit

sense resistor, R

. This pin is also the bias supply for the LED

current sense amplifier.

LN Negative sense amplifier connection for LED current limit

sense resistor, R

. The voltage at LN also determines whether

the boost or buck-boost mode is configured.

VREG Compensation capacitor for internal 5 V regulator.

LA Anode reference connection to LEDs. Using an external resis-

tor divider with the same ratio as the number of LEDs provides

a measurement of the voltage across all LEDs in the load. This

is compared to the voltage on the LF pin to provide shorted LED

detection. In addition, it is compared against voltage references to

Automotive High Current LED Controller

A6265

Allegro MicroSystems, LLC

115 Northeast Cutoff

Worcester, Massachusetts 01615-0036 U.S.A.

1.508.853.5000; www.allegromicro.com

provide open circuit or shorted LED string detection.

LF Single diode forward voltage reference input. Measures the

forward voltage of the first LED. This value is used as a reference

against the voltage on the LA pin to detect possible shorted LEDs

in the LED string.

Circuit Operation

Converter A constant frequency, current mode control scheme

is used to regulate the current through the LEDs. There are two

control loops within the regulator. The inner loop formed by the

amplifier, AS (see the Functional Block Diagram for AS, AC, AE,

and AL), comparator, AC, and the RS bistable, controls the induc-

tor current as measured through the switch by the switch sense

resistor, R

The outer loop including the amplifier, AL, and the integrating

error amplifier, AE, controls the average LED current by provid-

ing a setpoint reference for the inner loop.

The LED current is measured by the LED sense resistor, R

and compared to the internal reference current to produce an

integrated error signal at the output of AE. This error signal sets

the average amount of energy required from the inductor by the

LEDs. The average inductor energy transferred to the LEDs is

defined by the average inductor current as determined by the

inner control loop.

The inner loop establishes the average inductor current by

controlling the peak switch current on a cycle-by-cycle basis.

Because the relationship between peak current and average cur-

rent is non-linear, depending on the duty cycle, the reference

level for the peak switch current is modified by a slope generator.

This compensation reduces the peak switch current measurement

by a small amount as the duty cycle increases (refer to figure 1).

The slope compensation also removes the instability inherent in a

fixed frequency current control scheme.

The control loops work together as follows: the switch current,

sensed by the switch current sense resistor, R

, is compared

to the LED current error signal. As the LED current increases

the output of AE will reduce, reducing the peak switch current

and thus the current delivered to the LEDs. As the LED current

decreases the output of AE increases, increasing the peak switch

current and thus increasing the current delivered to the LEDs.

Under some conditions, especially when the LED current is set to

a low value, the energy required in the inductor may result in the

inductor current dropping to zero for part of each cycle. This is

known as discontinuous mode operation, and results in some low

frequency ripple. The average LED current, however, remains

regulated down to zero. In discontinuous mode, when the induc-

tor current drops to zero, the voltage at the drain of the external

MOSFET rings, due to the resonant LC circuit formed by the

inductor, and the switch and diode capacitance. This ringing is

low frequency and is not harmful.

Switch Current Limit The switch current is measured by the

switch sense resistor, R

, and the switch sense amplifier, AS

(see the Functional Block Diagram). The input limit of the sense

amplifier, V

IDS

, and the maximum switch current, I

SMAX

, define

the maximum value of the sense resistor as:

= V

IDS

/ I

SMAX (1)

This defines the maximum measurable value of the switch (and

inductor) current.

The maximum switch current is modulated by the on-time of the

switch. An internal slope compensation signal is subtracted from

the voltage sense signal to produce a peak sense voltage which

effectively defines the current limit. This signal is applied at a

rate of –50 mV / s starting with no contribution (t = 0 s) at the

beginning of each switching cycle. Figure 1 illustrates how the

peak sense voltage (typical values) changes over a period of 3 s.

For example, the maximum current (typical) through the switch

at t = 1.5 s (D = 50%) would be 410 mV/R

, however, if the

switch remained on for a further 1 s, the maximum current

through the switch would be 360 mV/R

500

450

400

350

300

250

200

150

100

Period (s)

Peak Sense Votlage (mV)

Figure 1. Slope compensation for peak switch current control.

Automotive High Current LED Controller

A6265

Allegro MicroSystems, LLC

115 Northeast Cutoff

Worcester, Massachusetts 01615-0036 U.S.A.

1.508.853.5000; www.allegromicro.com

LED Current Level The LED current is determined by a

combination of the LED sense resistor, R

, the LED current

threshold voltage, V

IDL

, and the voltage between the IREF pin

and GND ( V

IREF

The 100% current level, when the IREF pin is connected to

VREG, is defined as:

LED

(max) = V

IDL

/ R

(2)

If V

IREF

is less than 1 V then the 100% current level is defined as:

LED

(max) = V

IREF

/ (10 × R

) (3)

This feature provides direct analog dimming using a voltage from

0 to 1 V. This can be used to provide intensity-matching between

modules or groups of LEDs in critical display or backlighting

applications. It can also be used to provide a soft start, by con-

necting a capacitor from IREF to GND and a resistor from IREF

to VREG, or one-step dimming by use of a single logic control.

LED Brightness: PWM Dimming LED brightness can

be controlled by changing the current, which affects the light

intensity. However in some applications, for example with amber

LEDs, this will have some effect on the color of the LEDs.

In these cases it is more desirable to control the brightness by

switching the fixed LED current with a pulse width modulated

signal. This allows the LED brightness to be set with little effect

on the LED color and intensity and allows direct digital control

of the LED brightness.

A PWM signal can be applied to the EN input to enable PWM

dimming. The period of this signal should be less than the

minimum disable time, t

DIS

. During PWM dimming, the A6265

switches the LED current between 100% and typically 8% of

the full current. This ensures that the voltage change across the

LED string is limited to a few volts, depending on the number

of LEDs. This limits the stress on the load capacitor (across

the string of LEDs) due to large changes in voltage. If the load

capacitor is a multilayer ceramic type, then this will reduce any

audible noise due to the piezoelectric effect of the capacitor.

The rate of change of the LED current is also limited, to reduce

any large variations in the input current.

Sleep Mode If EN is held low for longer than the disable time,

DIS

, then the A6265 will shut down and put all sections into a

low-power sleep mode. In this mode the bias current is typically

less than 4

A. In the buck-boost configuration the only leakage

path remaining will be the path through the MOSFET.

Provided this is low, then the complete circuit may remain con-

nected to the power supply under all conditions. Note that the

disable time is derived from the oscillator period by a ratio of

32,768, so any variation in the oscillator frequency will change

the disable time.

Oscillator The main oscillator may be configured as a clock

source or it may be driven by an external clock signal. The oscil-

lator is designed to run between 100 and 700 kHz.

When the oscillator is configured as a clock source, the frequency

is controlled by a single external resistor, R

OSC

(k), between the

OSC pin and the GND pin. The oscillator frequency is approxi-

mately:

OSC

= 21700 / R

OSC

(kHz) (4)

Figure 2 shows the resulting f

OSC

for various values of R

OSC

If the OSC pin is connected to VREG or GND, the oscillator

frequency will be set internally to approximately 350 kHz.

When an external clock source is used to drive the OSC pin, it

can synchronize a number of A6265s operating together. This

ensures that only a single fundamental frequency is detectable

on the supply line, thus simplifying the design of any required

EMC filter. The disadvantage of using a single external clock

source is that all controllers will be switching current from the

supply at the same time. However, this effect may be reduced,

and the EMC performance may be further enhanced, by using

the CKOUT pin of another A6265 as the external clock source.

In this case the switching point of each subsequent A6265 in the

chain will be delayed from that of the previous A6265, and the

current pulses will be spread across the oscillator period.

700

600

500

400

300

200

100

30 50 70 11090 130 150 170 190 210

External Resistor Value, R

OSC

(k)

Oscillator Frequency, f

OSC

(kHz)

Figure 2. Internal oscillator frequency when set by R

OSC

P1-P3 P4-P6 P7-P9 P10-P12 P13-P15 P16-P17

A6265KLPTR-T

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