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LTC3863IDE#PBF

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LTC3863
13
3863fa
For more information www.linear.com/3863
The LTC3863 is a nonsynchronous inverting, current mode,
constant frequency PWM controller. It drives an external
P-channel power MOSFET which connects to a Schottky
power diode acting as the commutating catch diode. The
input range extends from 3.5V to 60V. The output range
has no theoretical minimum or maximum, but the duty
factor and external components practically limit the out
-
put to one-tenth and ten times the input voltage. Higher
output
ratios can be obtained with transformers and more
efficient external components.
The LTC3863 offers a highly efficient Burst Mode operation
with 70µA quiescent current, which delivers outstanding
efficiency in light load operation. The LTC3863 is a low
pin count, robust and easy-to-use inverting power supply
solution in applications which require high efficiency and
operate with widely varying input and output voltages.
The typical application on the front page is a basic LTC3863
application circuit. The LTC3863 can sense the inductor
current through a high side series sense resistor, R
SENSE
,
placed between V
IN
and the source of the external P-channel
MOSFET. Once the required output voltage and operating
frequency have been determined, external component
selection is driven by load requirements, and
begins with
the selection of inductor and R
SENSE
. Next, the power
MOSFET and catch diode are selected. Finally, input and
output capacitors are selected.
Output Voltage Programming
The output voltage is programmed by connecting a
feedback resistor divider from the output to the V
FB
pin
as shown in Figure 1. The output voltage in steady-state
operation is set by the feedback resistors according to
the equation:
V
OUT
= –0.8V •
R
FB1
R
FB2
applicaTions inForMaTion
Great care should be taken to route the V
FB
and V
FBN
lines away from noise sources, such as the inductor or
SW node or the GATE signal that drives the external P-
channel MOSFET.
The integrator capacitor, C
FB2
, should be sized to ensure
the negative sense amplifier gain rolls off and limits high
frequency gain peaking in the DC/DC control loop. The
integrator capacitor pole can be safely set to be two times
the switching frequency without affecting the DC/DC phase
margin according to the following equation. It is highly
recommended that C
FB2
be used in most applications.
C
FB2
=
1
2 • π • 2 • R
FB2
•FREQ
SW
The feedback resistor R
FB2
placed from the V
FBN
pin to
the V
FB
pin is driven from V
FB
by an internal amplifier with
limited V
FB
output current capability. For Operation at V
IN
down to 3.5V, it is recommended to limit the V
FB
output
current to 100µA, and R
FB2
should be no smaller than
8K. For V
IN
always greater than 6V, it is recommended to
limit the current to 250µA, and R
FB2
should be no smaller
than 3.2K. For minimum V
IN
between 3.5V and 6V, add
75µA for each volt V
IN-min
is above 3.5V to the maximum
current until reaching a maximum of 250µA at 6V
IN
, and
minimum R
FB2
would be 0.8V/lo-max.
LTC3863
V
FB
V
FBN
V
OUT
R
FB2
C
FB2
R
FB1
3863 F01
Figure 1. Setting the Output Voltage
LTC3863
14
3863fa
For more information www.linear.com/3863
applicaTions inForMaTion
Switching Frequency and Clock Synchronization
The choice of operating frequency is a trade-off between
efficiency and component size. Lowering the operating fre
-
quency improves efficiency by reducing MOSFET switching
losses but requires larger inductance and/or capacitance
to maintain low output ripple voltage. Conversely, raising
the operating frequency degrades efficiency but reduces
component size.
The LTC3863 can free-run at a user programmed switch
-
ing frequency, or it can synchronize with an external
clock to run at the clock frequency. When the LTC3863 is
synchronized, the GATE pin will synchronize in phase with
the rising edge of the applied clock in order to turn the
external P-channel MOSFET on. The switching frequency
of the LTC3863 is programmed with the FREQ pin, and the
external clock is applied at the PLLIN/MODE pin. Table 1
highlights the different states in which the FREQ pin can
be used in conjunction with the PLLIN/MODE pin.
Table 1
FREQ PIN PLLIN/MODE PIN FREQUENCY
OV DC Voltage 350kHz
Floating DC Voltage 535kHz
Resistor to GND DC Voltage 50kHz to 850kHz
Either of the Above External Clock Phase Locked to
External Clock
The free-running switching frequency can be programmed
from 50kHz to 850kHz by connecting a resistor from FREQ
pin to signal ground. The resulting switching frequency
as a function of resistance on the FREQ pin is shown in
Figure 2.
Set the free-running frequency to the desired synchroni
-
zation frequency using the FREQ pin so that the internal
oscillator is prebiased approximately to the synchronization
frequency. While it is not required that the free-running
frequency be near the external clock frequency, doing so
will minimize synchronization time.
Inductor Selection
Operating frequency, inductor selection, capacitor selection
and efficiency are interrelated. Higher operating frequen
-
cies allow the use of smaller inductors, smaller capacitors,
but
result in lower efficiency because of higher MOSFET
gate charge and transition losses. In addition to this basic
trade-off, the selection of inductor value is also influenced
by other factors.
Small inductor values result in large inductor ripple cur
-
rents, large output voltage ripples and low efficiency due
to higher core and conduction loss. Large inductor ripple
currents result
in high inductor peak currents, which re-
quire
physically
large inductors with large magnetic cross
sections and higher saturation current ratings.
The value of the inductor can also impact the stability of
the feedback loop. In continuous mode, the buck-boost
converter transfer function has a right-half plane zero at
a frequency that is inversely proportional to the value of
the inductor. As a result, large inductor values can move
this zero to a frequency that is low enough to degrade the
phase margin of the feedback loop. Large inductor values
also tend to degrade stability due to low noise margin
caused from low ripple current. Additionally, large value
inductors can lead to slow transient response due to slow
inductor current ramping time.
Figure 2. Switching Frequency vs Resistor on FREQ pin
FREQ PIN RESISTOR (kΩ)
15
FREQUENCY (kHz)
600
800
1000
35 45 5525
3863 F02
400
200
500
700
900
300
100
0
65 75 85 95 105 115
125
LTC3863
15
3863fa
For more information www.linear.com/3863
applicaTions inForMaTion
For an inverting buck-boost converter operating in con-
tinuous conduction
mode (CCM), given the desired input,
output voltages and switching frequency, the peak-to-peak
inductor ripple current is determined by the inductor value:
∆I
L(CCM)
=
V
IN
•D
L • f
=
V
IN
• |V
OUT
|+V
D
( )
L • f • V
IN
+|V
OUT
|+V
D
( )
where V
D
is the diode forward conduction voltage. In cases
where V
OUT
>> V
D
, V
D
can be ignored. D is the duty factor
and is given as:
D=
|V
OUT
|+V
D
V
IN
+|V
OUT
|+V
D
0 < D< 1
( )
The duty factor increases with increasing V
OUT
and de-
creasing V
IN
. For a given V
OUT
, the maximum duty factor
occurs at minimum V
IN
.
A typical starting point for selecting an inductor is to choose
the inductance such that the maximum peak-to-peak in-
ductor ripple current, ∆
I
L(MAX)
, is set to 40% ~ 50% of the
inductor average current, I
L(AVG)
, at maximum load current.
Since ∆I
L(MAX)
occurs at maximum V
IN
in continuous mode,
the inductance is calculated at maximum V
IN
:
L =
V
IN(MAX)
2
• |V
OUT
|+V
D
( )
0.4•I
OUT(MAX)
• f • V
IN(MAX)
+|V
OUT
|+V
D
( )
2
The inductance can be further adjusted to achieve specific
design optimization of efficiency, output ripple, component
size and loop response.
Once the inductance value has been determined, the type
of inductor must be selected. Core loss is independent of
core size for a given inductor value, but it is very depen
-
dent on the inductance selected. As inductance increases,
core
losses decrease. Unfortunately, increased inductance
requires more turns of wire and therefore, copper losses
will increase.
High efficiency converters generally cannot tolerate the
core loss of low cost powdered iron cores, forcing the use
of more expensive ferrite materials. Ferrite designs have
very low core loss and are preferred at high switching
frequencies, so design goals can concentrate on cop
-
per loss and preventing saturation. Ferrite core material
saturates
hard, which means that inductance collapses
abruptly when the peak design current is exceeded. This
will result in an abrupt increase in inductor ripple current
and output voltage ripple. Do not allow the core to saturate!
A variety of inductors are available from manufacturers
such as Sumida, Panasonic, Coiltronics, Coilcraft, Toko,
Vishay, Pulse and Würth.
Current Sensing and Current Limit Programming
The LTC3863 senses the inductor current through a cur
-
rent sense resistor, R
SENSE
, placed across the V
IN
and
SENSE pins. The voltage across the resistor, V
SENSE
, is
proportional to inductor current and in normal operation is
compared to the peak inductor current setpoint. An inductor
current limit condition is detected when V
SENSE
exceeds
95mV. When the current limit threshold is exceeded, the
P-channel MOSFET is immediately turned off by pulling
the GATE voltage to V
IN
regardless of the controller input.
The peak inductor current limit is equal to:
I
L(PEAK)
≅
95mV
R
SENSE
This inductor current limit would translate to an output
current limit based on the inductor ripple and duty factor:
I
OUT(LIMIT)
=
95mV
R
SENSE
–
∆I
L
2
• 1–D
( )
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LTC3863IDE#PBF

Mfr. #:
Buy LTC3863IDE#PBF
Manufacturer:
Analog Devices / Linear Technology
Description:
Switching Voltage Regulators 60V Low IQ Inverting Buck-Boost DC/DC PMOS Controller
Lifecycle:
New from this manufacturer.
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