LTC3728LEGN-1#TRPBF Datasheet LTC3728LIGN-1#PBF, LTC3728LIUH-1#PBF, LTC3728LEGN-1#TRPBF | P22-P24

LTC3728L-1

3728l1fc

APPLICATIONS INFORMATION

If V

SEC

drops below this level, the FCB voltage forces

temporary continuous switching operation until V

SEC

again above its minimum.

In order to prevent erratic operation if no external connec-

tions are made to the FCB pin, the FCB pin has a 0.18µA

internal current source pulling the pin high. Include this

current when choosing resistor values R5 and R6.

The following table summarizes the possible states avail-

able on the FCB pin:

Table 1

FCB PIN CONDITION

0V to 0.75V Forced Continuous Both Controllers

(Current Reversal Allowed—

Burst Inhibited)

0.85V < V

FCB

< 4.3V Minimum Peak Current Induces

Burst Mode Operation

No Current Reversal Allowed

Feedback Resistors Regulating a Secondary Winding

>4.8V Burst Mode Operation Disabled

Constant Frequency Mode Enabled

No Current Reversal Allowed

No Minimum Peak Current

Voltage Positioning

Voltage positioning can be used to minimize peak-to-peak

output voltage excursions under worst-case transient

loading conditions. The open-loop DC gain of the control

loop is reduced depending upon the maximum load step

speciﬁ cations. Voltage positioning can easily be added

to either or both controllers by loading the I

pin with

a resistive divider having a Thevenin equivalent voltage

source equal to the midpoint operating voltage range of

the error ampliﬁ er, or 1.2V (see Figure 8).

The resistive load reduces the DC loop gain while main-

taining the linear control range of the error ampliﬁ er.

The maximum output voltage deviation can theoretically

be reduced to half or alternatively the amount of output

capacitance can be reduced for a particular application. A

complete explanation is included in Design Solutions 10.

(See www.linear.com)

Efﬁ ciency Considerations

The percent efﬁ ciency of a switching regulator is equal to

the output power divided by the input power times 100%.

It is often useful to analyze individual losses to determine

what is limiting the efﬁ ciency and which change would

produce the most improvement. Percent efﬁ ciency can

be expressed as:

%Efﬁ ciency = 100% – (L1 + L2 + L3 + ...)

where L1, L2, etc. are the individual losses as a percent-

age of input power.

Although all dissipative elements in the circuit produce

losses, four main sources usually account for most of the

losses in LTC3728L-1 circuits: 1) IC V

current (including

loading on the 3.3V internal regulator), 2) INTV

regula-

tor current, 3) I

R losses, 4) Topside MOSFET transition

losses.

1. The V

current has two components: the ﬁ rst is the DC

supply current given in the Electrical Characteristics table,

which excludes MOSFET driver and control currents; the

second is the current drawn from the 3.3V linear regulator

output. V

current typically results in a small (<0.1%)

loss.

Figure 8. Active Voltage Positioning

Applied to the LTC3728L-1

INTV

3728L1 F08

LTC3728L-1

3728l1fc

APPLICATIONS INFORMATION

2. INTV

current is the sum of the MOSFET driver and

control currents. The MOSFET driver current results from

switching the gate capacitance of the power MOSFETs.

Each time a MOSFET gate is switched from low to high

to low again, a packet of charge dQ moves from INTV

to ground. The resulting dQ/dt is a current out of INTV

that is typically much larger than the control circuit cur-

rent. In continuous mode, I

GATECHG

=f(Q

), where Q

and Q

are the gate charges of the topside and bottom

side MOSFETs.

Supplying INTV

power through the EXTV

switch input

from an output-derived source will scale the V

current

required for the driver and control circuits by a factor of

(Duty Cycle)/(Efﬁ ciency). For example, in a 20V to 5V ap-

plication, 10mA of INTV

current results in approximately

2.5mA of V

current. This reduces the mid-current loss

from 10% or more (if the driver was powered directly from

) to only a few percent.

3. I

R losses are predicted from the DC resistances of

the fuse (if used), MOSFET, inductor, current sense resis-

tor, and input and output capacitor ESR. In continuous

mode the average output current ﬂ ows through L and

SENSE

, but is “chopped” between the topside MOSFET

and the synchronous MOSFET. If the two MOSFETs have

approximately the same R

DS(ON)

, then the resistance of

one MOSFET can simply be summed with the resistances

of L, R

SENSE

and ESR to obtain I

R losses. For example, if

each R

DS(ON)

= 30m, R

= 50m, R

SENSE

= 10m and

ESR

= 40m (sum of both input and output capacitance

losses), then the total resistance is 130m. This results

in losses ranging from 3% to 13% as the output current

increases from 1A to 5A for a 5V output, or a 4% to 20%

loss for a 3.3V output. Efﬁ ciency varies as the inverse

square of V

OUT

for the same external components and

output power level. The combined effects of increasingly

lower output voltages and higher currents required by

high performance digital systems is not doubling but

quadrupling the importance of loss terms in the switching

regulator system!

4. Transition losses apply only to the topside MOSFET(s),

and become signiﬁ cant only when operating at high input

voltages (typically 15V or greater). Transition losses can

be estimated from:

Transition Loss = V

()

•

MAX













()

•

MILLER

()

5V – V













Other “hidden” losses such as copper trace and internal

battery resistances can account for an additional 5% to

10% efﬁ ciency degradation in portable systems. It is very

important to include these “system” level losses during

the design phase. The internal battery and fuse resistance

losses can be minimized by making sure that C

has ad-

equate charge storage and very low ESR at the switching

frequency. A 25W supply will typically require a minimum

of 20µF to 40µF of capacitance having a maximum of 20m

to 50m of ESR. The LTC3728L-1 2-phase architecture

typically halves this input capacitance requirement over

competing solutions. Other losses including Schottky con-

duction losses during dead-time and inductor core losses

generally account for less than 2% total additional loss.

Checking Transient Response

The regulator loop response can be checked by looking at

the load current transient response. Switching regulators

take several cycles to respond to a step in DC (resistive)

load current. When a load step occurs, V

OUT

shifts by

an amount equal to ∆I

LOAD

(ESR), where ESR is the ef-

fective series resistance of C

OUT

. ∆I

LOAD

also begins to

charge or discharge C

OUT

generating the feedback error

signal that forces the regulator to adapt to the current

change and return V

OUT

to its steady-state value. During

this recovery time V

OUT

can be monitored for excessive

overshoot or ringing, which would indicate a stability

problem. OPTI-LOOP compensation allows the transient

response to be optimized over a wide range of output

capacitance and ESR values.

The availability of the I

not only allows optimization of control loop behavior but

also provides a DC coupled and AC ﬁ ltered closed loop

response test point. The DC step, rise time and settling

LTC3728L-1

3728l1fc

APPLICATIONS INFORMATION

at this test point truly reﬂ ects the closed loop response.

Assuming a predominantly second order system, phase

margin and/or damping factor can be estimated using the

percentage of overshoot seen at this pin. The bandwidth

can also be estimated by examining the rise time at the

pin. The I

external components shown in the Figure 1

circuit will provide an adequate starting point for most

applications.

The I

series R

-C

ﬁ lter sets the dominant pole-zero

loop compensation. The values can be modiﬁ ed slightly

(from 0.5 to 2 times their suggested values) to optimize

transient response once the ﬁ nal PC layout is done and

the particular output capacitor type and value have been

determined. The output capacitors need to be selected

because the various types and values determine the loop

gain and phase. An output current pulse of 20% to 80%

of full-load current having a rise time of 1µs to 10µs will

produce output voltage and I

pin waveforms that will

give a sense of the overall loop stability without break-

ing the feedback loop. Placing a power MOSFET directly

across the output capacitor and driving the gate with an

appropriate signal generator is a practical way to produce

a realistic load step condition. The initial output voltage

step resulting from the step change in output current may

not be within the bandwidth of the feedback loop, so this

signal cannot be used to determine phase margin. This

is why it is better to look at the I

pin signal which is in

the feedback loop and is the ﬁ ltered and compensated

control loop response. The gain of the loop will be in-

creased by increasing R

and the bandwidth of the loop

will be increased by decreasing C

. If R

is increased by

the same factor that C

is decreased, the zero frequency

will be kept the same, thereby keeping the phase shift the

same in the most critical frequency range of the feedback

loop. The output voltage settling behavior is related to the

stability of the closed-loop system and will demonstrate

the actual overall supply performance.

A second, more severe transient is caused by switching

in loads with large (>1µF) supply bypass capacitors. The

discharged bypass capacitors are effectively put in parallel

with C

OUT

, causing a rapid drop in V

OUT

. No regulator can

alter its delivery of current quickly enough to prevent this

sudden step change in output voltage if the load switch

resistance is low and it is driven quickly. If the ratio of

LOAD

to C

OUT

is greater than 1:50, the switch rise time

should be controlled so that the load rise time is limited

to approximately 25 • C

LOAD

. Thus a 10µF capacitor would

require a 250µs rise time, limiting the charging current

to about 200mA.

Automotive Considerations: Plugging into the

Cigarette Lighter

As battery-powered devices go mobile, there is a natural

interest in plugging into the cigarette lighter in order to

conserve or even recharge battery packs during opera-

tion. But before you connect, be advised: you are plug-

ging into the supply from hell. The main power line in an

automobile is the source of a number of nasty potential

transients, including load-dump, reverse-battery, and

double-battery.

Load-dump is the result of a loose battery cable. When the

cable breaks connection, the ﬁ eld collapse in the alterna-

tor can cause a positive spike as high as 60V which takes

several hundred milliseconds to decay. Reverse-battery is

just what it says, while double-battery is a consequence of

tow-truck operators ﬁ nding that a 24V jump start cranks

cold engines faster than 12V.

The network shown in Figure 9 is the most straightforward

approach to protect a DC/DC converter from the ravages

of an automotive power line. The series diode prevents

current from ﬂ owing during reverse-battery, while the

transient suppressor clamps the input voltage during

load-dump. Note that the transient suppressor should not

conduct during double-battery operation, but must still

clamp the input voltage below breakdown of the converter.

Although the LTC3728L-1 has a maximum input voltage

of 30V, most applications will also be limited to 30V by

the MOSFET BVD

Figure 9. Automotive Application Protection

3728L1 F09

LTC3728L-1

TRANSIENT VOLTAGE

SUPPRESSOR

GENERAL INSTRUMENT

1.5KA24A

50A I

RATING

12V

P1-P3 P4-P6 P7-P9 P10-P12 P13-P15 P16-P18 P19-P21 P22-P24 P25-P27 P28-P30 P31-P32

LTC3728LEGN-1#TRPBF

Mfr. #:

Buy LTC3728LEGN-1#TRPBF

Manufacturer:

Analog Devices / Linear Technology

Description:

Switching Voltage Regulators 2x, 550kHz, 2-PhSync Reg

Lifecycle:

New from this manufacturer.

Delivery:

DHL FedEx Ups TNT EMS

Payment:

T/T Paypal Visa MoneyGram Western Union

LTC3728LEGN-1#TRPBF

LTC3728LEGN-1#TRPBF

Products related to this Datasheet