22
LTC1709-7
synchronous MOSFET. If the two MOSFETs have approxi-
mately the same R
DS(ON)
, then the resistance of one
MOSFET can simply be summed with the resistances of L,
R
SENSE
and ESR to obtain I
2
R losses. For example, if each
R
DS(ON)
= 10mΩ, R
L
= 10mΩ, and R
SENSE
= 5mΩ, then the
total resistance is 25mΩ. This results in losses ranging
from 2% to 8% as the output current increases from 3A to
15A per output stage for a 5V output, or a 3% to 12% loss
per output stage for a 3.3V output. Efficiency varies as the
inverse square of V
OUT
for the same external components
and output power level. The combined effects of increas-
ingly lower output voltages and higher currents required
by high performance digital systems is not doubling but
quadrupling the importance of loss terms in the switching
regulator system!
2) Transition losses apply only to the topside MOSFET(s),
and are significant only when operating at high input
voltages (typically 12V or greater). Transition losses can
be estimated from:
Transition Loss = (1.7) V
IN
2
I
O(MAX)
C
RSS
f
3) INTV
CC
current is the sum of the MOSFET driver and
control currents. The MOSFET driver current results from
switching the gate capacitance of the power MOSFETs.
Each time a MOSFET gate is switched from low to high to
low again, a packet of charge dQ moves from INTV
CC
to
ground. The resulting dQ/dt is a current out of INTV
CC
that
is typically much larger than the control circuit current. In
continuous mode, I
GATECHG
= (Q
T
+ Q
B
), where Q
T
and Q
B
are the gate charges of the topside and bottom side
MOSFETs.
Supplying INTV
CC
power through the EXTV
CC
switch input
from an output-derived source will scale the V
IN
current
required for the driver and control circuits by the ratio
(Duty Factor)/(Efficiency). For example, in a 20V to 5V
application, 10mA of INTV
CC
current results in approxi-
mately 3mA of V
IN
current. This reduces the mid-current
loss from 10% or more (if the driver was powered directly
from V
IN
) to only a few percent.
4) The V
IN
current has two components: the first is the
DC supply current given in the Electrical Characteristics
table, which excludes MOSFET driver and control cur-
rents; the second is the current drawn from the differential
amplifier output. V
IN
current typically results in a small
(<0.1%) loss.
Other “hidden” losses such as copper trace and internal
battery resistances can account for an additional 5% to
10% efficiency degradation in portable systems. It is very
important to include these “system” level losses in the
design of a system. The internal battery and input fuse
resistance losses can be minimized by making sure that
C
IN
has adequate charge storage and a very low ESR at
the switching frequency. A 50W supply will typically
require a minimum of 200µF to 300µF of output capaci-
tance having a maximum of 10mΩ to 20mΩ of ESR. The
LTC1709-7 2-phase architecture typically halves the
input and output capacitance requirements over compet-
ing solutions. Other losses including Schottky conduc-
tion losses during dead-time and inductor core losses
generally account for less than 2% total additional loss.
Checking Transient Response
The regulator loop response can be checked by looking at
the load transient response. Switching regulators take
several cycles to respond to a step in DC (resistive) load
current. When a load step occurs, V
OUT
shifts by an
amount equal to ∆I
LOAD
(ESR), where ESR is the effective
series resistance of C
OUT
(∆I
LOAD
) also begins to charge or
discharge C
OUT
generating the feedback error signal that
forces the regulator to adapt to the current change and
return V
OUT
to its steady-state value. During this recovery
time V
OUT
can be monitored for excessive overshoot or
ringing, which would indicate a stability problem.
The
availability of the I
TH
pin not only allows optimization of
control loop behavior but also provides a DC coupled and
AC filtered closed loop response test point. The DC step,
rise time, and settling at this test point truly reflects the
closed loop response.
Assuming a predominantly second
order system, phase margin and/or damping factor can be
estimated using the percentage of overshoot seen at this
pin. The bandwidth can also be estimated by examining
the rise time at the pin. The I
TH
external components
shown in the Figure 1 circuit will provide an adequate
starting point for most applications.
The I
TH
series R
C
-C
C
filter sets the dominant pole-zero
loop compensation. The values can be modified slightly
APPLICATIO S I FOR ATIO
WUU
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