LTC3727AIG-1#PBF Datasheet LTC3727AIG-1#PBF, LTC3727AEG-1#TRPBF, LTC3727AIG-1#TRPBF | P10-P12

LTC3727A-1

3727a1fa

Main Control Loop

The LTC3727A-1 uses a constant frequency, current

mode step-down architecture with the two controller

channels operating 180 degrees out of phase. During

normal operation, each top MOSFET is turned on when

the clock for that channel sets the RS latch, and turned

off when the main current comparator, I

, resets the RS

latch. The peak inductor current at which I

resets the RS

latch is controlled by the voltage on the I

pin, which is

the output of each error ampliﬁ er EA. The V

OSENSE

receives the voltage feedback signal, which is compared

to the internal reference voltage by the EA. When the load

current increases, it causes a slight decrease in V

OSENSE

relative to the 0.8V reference, which in turn causes the

voltage to increase until the average inductor current

matches the new load current. After the top MOSFET has

turned off, the bottom MOSFET is turned on until either the

inductor current starts to reverse, as indicated by current

comparator I

, or the beginning of the next cycle.

The top MOSFET drivers are biased from ﬂ oating bootstrap

capacitor C

, which normally is recharged during each off

cycle through an external diode when the top MOSFET

turns off. As V

decreases to a voltage close to V

OUT

the loop may enter dropout and attempt to turn on the

top MOSFET continuously. The dropout detector detects

this and forces the top MOSFET off for about 400ns every

tenth cycle to allow C

to recharge.

The main control loop is shut down by pulling the

RUN/SS pin low. Releasing RUN/SS allows an internal

1.2μA current source to charge soft-start capacitor C

When C

reaches 1.5V, the main control loop is enabled

with the I

voltage clamped at approximately 30% of its

maximum value. As C

continues to charge, the I

voltage is gradually released allowing normal, full-current

operation. When both RUN/SS1 and RUN/SS2 are low, all

LTC3727A-1 controller functions are shut down, including

the 7.5V and 3.3V regulators.

Low Current Operation

The FCB pin is a multifunction pin providing two func-

tions: 1) to provide regulation for a secondary winding

by temporarily forcing continuous PWM operation on

both controllers; and 2) to select between two modes of

low current operation. When the FCB pin voltage is below

0.8V, the controller forces continuous PWM current mode

operation. In this mode, the top and bottom MOSFETs are

alternately turned on to maintain the output voltage inde-

pendent of direction of inductor current. When the FCB pin

is below V

INTVCC

– 2V but greater than 0.8V, the controller

enters Burst Mode operation. Burst Mode operation sets

a minimum output current level before inhibiting the top

switch and turns off the synchronous MOSFET(s) when

the inductor current goes negative. This combination of

requirements will, at low currents, force the I

pin below

a voltage threshold that will temporarily inhibit turn-on of

both output MOSFETs until the output voltage drops. There

is 60mV of hysteresis in the burst comparator B tied to

the I

pin. This hysteresis produces output signals to the

MOSFETs that turn them on for several cycles, followed by

a variable “sleep” interval depending upon the load cur-

rent. The resultant output voltage ripple is held to a very

small value by having the hysteretic comparator follow

the error ampliﬁ er gain block.

Frequency Synchronization

The phase-locked loop allows the internal oscillator to

be synchronized to an external source via the PLLIN pin.

The output of the phase detector at the PLLFLTR pin is

also the DC frequency control input of the oscillator that

operates over a 250kHz to 550kHz range corresponding

to a DC voltage input from 0V to 2.4V. When locked, the

PLL aligns the turn on of the top MOSFET to the rising

edge of the synchronizing signal. When PLLIN is left

open, the PLLFLTR pin goes low, forcing the oscillator to

its minimum frequency.

OPERATION

(Refer to Functional Diagram)

LTC3727A-1

3727a1fa

Continuous Current (PWM) Operation

Tying the FCB pin to ground will force continuous current

operation. This is the least efﬁ cient operating mode, but

may be desirable in certain applications. The output can

source or sink current in this mode. When sinking cur-

rent while in forced continuous operation, current will be

forced back into the main power supply.

INTV

/EXTV

Power

Power for the top and bottom MOSFET drivers and most

other internal circuitry is derived from the INTV

pin.

When the EXTV

pin is left open, an internal 7.5V low

dropout linear regulator supplies INTV

power. If EXTV

is taken above 7.3V, the 7.5V regulator is turned off and

an internal switch is turned on connecting EXTV

INTV

. This allows the INTV

power to be derived from

a high efﬁ ciency external source such as the output of the

regulator itself or a secondary winding, as described in

the Applications Information section.

Output Overvoltage Protection

An overvoltage comparator, OV, guards against transient

overshoots (>7.5%) as well as other more serious

conditions that may overvoltage the output. In this case,

the top MOSFET is turned off and the bottom MOSFET is

turned on until the overvoltage condition is cleared.

Power Good (PGOOD) Pin

The PGOOD pin is connected to an open drain of an internal

MOSFET. The MOSFET turns on and pulls the pin low when

either output is not within ±7.5% of the nominal output

level as determined by the resistive feedback divider. When

both outputs meet the ±7.5% requirement, the MOSFET is

turned off within 10μs and the pin is allowed to be pulled

up by an external resistor to a source of up to 7V.

Theory and Beneﬁ ts of 2-Phase Operation

The LTC3727A-1 dual high efﬁ ciency DC/DC controller

brings the considerable beneﬁ ts of 2-phase operation

to portable applications. Notebook computers, PDAs,

handheld terminals and automotive electronics will all

beneﬁ t from the lower input ﬁ ltering requirement, reduced

electromagnetic interference (EMI) and increased efﬁ ciency

associated with 2-phase operation.

Traditionally, constant-frequency dual switching regula-

tors operated both channels in phase (i.e., single-phase

operation). This means that both switches turned on at

the same time, causing current pulses of up to twice the

amplitude of those for one regulator to be drawn from the

input capacitor and battery. These large amplitude current

pulses increased the total RMS current ﬂ owing from the

input capacitor, requiring the use of more expensive input

capacitors and increasing both EMI and losses in the input

capacitor and battery.

With 2-phase operation, the two channels of the

dual-switching regulator are operated 180 degrees out of

phase. This effectively interleaves the current pulses drawn

by the switches, greatly reducing the overlap time where

they add together. The result is a signiﬁ cant reduction in

total RMS input current, which in turn allows less expensive

input capacitors to be used, reduces shielding requirements

for EMI and improves real world operating efﬁ ciency.

Figure 3 compares the input waveforms for a representative

single-phase dual switching regulator to the LTC3727A-1

2-phase dual switching regulator. An actual measure-

ment of the RMS input current under these conditions

shows that 2-phase operation dropped the input current

from 2.53A

RMS

to 1.55A

RMS

. While this is an impressive

reduction in itself, remember that the power losses are

proportional to I

RMS

, meaning that the actual power wasted

OPERATION

(Refer to Functional Diagram)

LTC3727A-1

3727a1fa

OPERATION

is reduced by a factor of 2.66. The reduced input ripple

voltage also means less power is lost in the input power

path, which could include batteries, switches, trace/con-

nector resistances and protection circuitry. Improvements

in both conducted and radiated EMI also directly accrue as

a result of the reduced RMS input current and voltage.

Of course, the improvement afforded by 2-phase opera-

tion is a function of the dual switching regulator’s relative

duty cycles which, in turn, are dependent upon the input

voltage V

(Duty Cycle = V

OUT

). Figure 4 shows how

the RMS input current varies for single-phase and 2-phase

operation for 3.3V and 5V regulators over a wide input

voltage range.

It can readily be seen that the advantages of 2-phase opera-

tion are not just limited to a narrow operating range, but

in fact extend over a wide region. A good rule of thumb

for most applications is that 2-phase operation will reduce

the input capacitor requirement to that for just one channel

operating at maximum current and 50% duty cycle.

5V SWITCH

20V/DIV

3.3V SWITCH

20V/DIV

INPUT CURRENT

5A/DIV

INPUT VOLTAGE

500mV/DIV

IN(MEAS)

= 2.53A

RMS

3727 G03a

(a)

IN(MEAS)

= 1.55A

RMS

3727 G03b

(b)

Figure 3. Input Waveforms Comparing Single-Phase (a) and 2-Phase (b) Operation for

Dual Switching Regulators Converting 12V to 5V and 3.3V at 3A Each. The Reduced Input

Ripple with the LTC3727A-1 2-Phase Regulator Allows Less Expensive Input Capacitors,

Reduces Shielding Requirements for EMI and Improves Efﬁ ciency

Figure 4. RMS Input Current Comparison

INPUT VOLTAGE (V)

INPUT RMS CURRENT (A)

3.0

2.5

2.0

1.5

1.0

0.5

10 20 30 40

3727 F04

SINGLE PHASE

DUAL CONTROLLER

2-PHASE

DUAL CONTROLLER

= 5V/3A

= 3.3V/3A

(Refer to Functional Diagram)

P1-P3 P4-P6 P7-P9 P10-P12 P13-P15 P16-P18 P19-P21 P22-P24 P25-P27 P28-P30 P31-P32

LTC3727AIG-1#PBF

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Manufacturer:

Analog Devices / Linear Technology

Description:

Switching Voltage Regulators Dual, 2-Phase Synchronous Controller w/ up to 14V Output

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