OMO ElectronicOMO Electronic
Expand menu
Hello, Sign inMy Account
0 Cart

LTC3727AIG-1#PBF

P1-P3P4-P6P7-P9P10-P12P13-P15P16-P18P19-P21P22-P24P25-P27P28-P30P31-P32
LTC3727A-1
22
3727a1fa
Effi ciency Considerations
The percent effi ciency of a switching regulator is equal to
the output power divided by the input power times 100%.
It is often useful to analyze individual losses to determine
what is limiting the effi ciency and which change would
produce the most improvement. Percent effi ciency can
be expressed as:
%Effi ciency = 100% – (L1 + L2 + L3 + ...)
where L1, L2, etc. are the individual losses as a percentage
of input power.
Although all dissipative elements in the circuit produce
losses, four main sources usually account for most of
the losses in LTC3727A-1 circuits: 1) LTC3727A-1 V
IN
current (including loading on the 3.3V internal regulator),
2) INTV
CC
regulator current, 3) I
2
R losses, 4) Topside
MOSFET transition losses.
1. The V
IN
current has two components: the fi rst is the
DC supply current given in the Electrical Characteris-
tics table, which excludes MOSFET driver and control
currents; the second is the current drawn from the 3.3V
linear regulator output. V
IN
current typically results in
a small (<0.1%) loss.
2. INTV
CC
current is the sum of the MOSFET driver and
control currents. The MOSFET driver current results from
switching the gate capacitance of the power MOSFETs.
Each time a MOSFET gate is switched from low to high to
low again, a packet of charge dQ moves from INTV
CC
to
ground. The resulting dQ/dt is a current out of INTV
CC
that
is typically much larger than the control circuit current.
In continuous mode, I
GATECHG
=f(Q
T
+ Q
B
), where Q
T
and
Q
B
are the gate charges of the topside and bottom side
MOSFETs.
Supplying INTV
CC
power through the EXTV
CC
switch input
from an output-derived source will scale the V
IN
current
required for the driver and control circuits by a factor of
(Duty Cycle)/(Effi ciency). For example, in a 20V to 5V
application, 10mA of INTV
CC
current results in approxi-
mately 2.5mA of V
IN
current. This reduces the mid-current
loss from 10% or more (if the driver was powered directly
from V
IN
) to only a few percent.
APPLICATIONS INFORMATION
3. I
2
R losses are predicted from the DC resistances of
the fuse (if used), MOSFET, inductor, current sense resis-
tor, and input and output capacitor ESR. In continuous
mode the average output current fl ows through L and
R
SENSE
, but is “chopped” between the topside MOSFET
and the synchronous MOSFET. If the two MOSFETs have
approximately the same R
DS(ON)
, then the resistance of
one MOSFET can simply be summed with the resistances
of L, R
SENSE
and ESR to obtain I
2
R losses. For example, if
each R
DS(ON)
= 30mΩ, R
L
= 50mΩ, R
SENSE
= 10mΩ and
R
ESR
= 40mΩ (sum of both input and output capacitance
losses), then the total resistance is 130mΩ. This results
in losses ranging from 3% to 13% as the output current
increases from 1A to 5A for a 5V output, or a 4% to 20%
loss for a 3.3V output. Effi ciency varies as the inverse
square of V
OUT
for the same external components and
output power level. The combined effects of increasingly
lower output voltages and higher currents required by
high performance digital systems is not doubling but
quadrupling the importance of loss terms in the switching
regulator system!
Figure 8. Active Voltage Positioning Applied to the LTC3727A-1
I
TH
R
C
R
T1
INTV
CC
C
C
3727 F08
LTC3727A-1
R
T2
4. Transition losses apply only to the topside MOSFET(s),
and become signifi cant only when operating at high input
voltages (typically 15V or greater). Transition losses can
be estimated from:
Transition Loss = (1.7) V
IN
2
I
O(MAX)
C
RSS
f
Other “hidden” losses such as copper trace and internal
battery resistances can account for an additional 5% to
10% effi ciency degradation in portable systems. It is very
important to include these “system” level losses during
the design phase. The internal battery and fuse resistance
losses can be minimized by making sure that C
IN
has
LTC3727A-1
23
3727a1fa
adequate charge storage and very low ESR at the switching
frequency. A 25W supply will typically require a minimum
of 22μF to 47μF of capacitance having a maximum of 20mΩ
to 50mΩ of ESR. The LTC3727A-1 2-phase architecture
typically halves this input capacitance requirement over
competing solutions. Other losses, including Schottky
diode conduction losses during dead-time and inductor
core losses, generally account for less than 2% total
additional loss.
Checking Transient Response
The regulator loop response can be checked by looking at
the load current transient response. Switching regulators
take several cycles to respond to a step in DC (resistive)
load current. When a load step occurs, V
OUT
shifts by an
amount equal to ΔI
LOAD
(ESR), where ESR is the effective
series resistance of C
OUT
. ΔI
LOAD
also begins to charge or
discharge C
OUT
generating the feedback error signal that
forces the regulator to adapt to the current change and
return V
OUT
to its steady-state value. During this recov-
ery time V
OUT
can be monitored for excessive overshoot
or ringing, which would indicate a stability problem.
OPTI-LOOP compensation allows the transient response
to be optimized over a wide range of output capacitance
and ESR values. The availability of the I
TH
pin not only
allows optimization of control loop behavior but also pro-
vides a DC coupled and AC fi ltered closed loop response
test point. The DC step, rise time and settling at this test
point truly refl ects the closed loop response. Assuming a
predominantly second order system, phase margin and/or
damping factor can be estimated using the percentage of
overshoot seen at this pin. The bandwidth can also be esti-
mated by examining the rise time at the pin. The I
TH
external
components shown in the Figure 1 circuit will provide an
adequate starting point for most applications.
The I
TH
series R
C
-C
C
fi lter sets the dominant pole-zero
loop compensation. The values can be modifi ed slightly
(from 0.5 to 2 times their suggested values) to optimize
transient response once the fi nal PC layout is done and
the particular output capacitor type and value have been
determined. The output capacitors need to be selected
because the various types and values determine the loop
APPLICATIONS INFORMATION
gain and phase. An output current pulse of 20% to 80%
of full-load current having a rise time of 1μs to 10μs will
produce output voltage and I
TH
pin waveforms that will
give a sense of the overall loop stability without breaking
the feedback loop. Placing a power MOSFET directly
across the output capacitor and driving the gate with an
appropriate signal generator is a practical way to produce
a realistic load step condition. The initial output voltage
step resulting from the step change in output current may
not be within the bandwidth of the feedback loop, so this
signal cannot be used to determine phase margin. This is
why it is better to look at the I
TH
pin signal which is in the
feedback loop and is the fi ltered and compensated control
loop response. The gain of the loop will be increased
by increasing R
C
and the bandwidth of the loop will be
increased by decreasing C
C
. If R
C
is increased by the same
factor that C
C
is decreased, the zero frequency will be kept
the same, thereby keeping the phase shift the same in the
most critical frequency range of the feedback loop. The
output voltage settling behavior is related to the stability
of the closed-loop system and will demonstrate the actual
overall supply performance.
A second, more severe transient is caused by switching
in loads with large (>1μF) supply bypass capacitors. The
discharged bypass capacitors are effectively put in parallel
with C
OUT
, causing a rapid drop in V
OUT
. No regulator can
alter its delivery of current quickly enough to prevent this
sudden step change in output voltage if the load switch
resistance is low and it is driven quickly. If the ratio of
C
LOAD
to C
OUT
is greater than 1:50, the switch rise time
should be controlled so that the load rise time is limited
to approximately 25 • C
LOAD
. Thus a 10μF capacitor would
require a 250μs rise time, limiting the charging current
to about 200mA.
Automotive Considerations: Plugging into the
Cigarette Lighter
As battery-powered devices go mobile, there is a natural
interest in plugging into the cigarette lighter in order to
conserve or even recharge battery packs during operation.
But before you connect, be advised: you are plugging
into the supply from Hell. The main power line in an
LTC3727A-1
24
3727a1fa
automobile is the source of a number of nasty potential
transients, including load-dump, reverse-battery, and
double-battery.
Load-dump is the result of a loose battery cable. When the
cable breaks connection, the fi eld collapse in the alterna-
tor can cause a positive spike as high as 60V which takes
several hundred milliseconds to decay. Reverse-battery is
just what it says, while double-battery is a consequence of
tow-truck operators fi nding that a 24V jump start cranks
cold engines faster than 12V.
The network shown in Figure 9 is the most straight for-
ward approach to protect a DC/DC converter from the
ravages of an automotive power line. The series diode
prevents current from fl owing during reverse-battery,
while the transient suppressor clamps the input voltage
during load-dump. Note that the transient suppressor
should not conduct during double-battery operation, but
must still clamp the input voltage below breakdown of
the converter. Although the LTC3727A-1 has a maximum
input voltage of 36V, most applications will be limited to
30V by the MOSFET BVDSS.
Design Example
As a design example for one channel, assume V
IN
=
24V(nominal), V
IN
= 30V(max), V
OUT
= 12V, I
MAX
= 5A
and f = 250kHz.
The inductance value is chosen fi rst based on a 40% ripple
current assumption. The highest value of ripple current
occurs at the maximum input voltage. Tie the PLLFLTR
pin to the SGND pin for 250kHz operation. The minimum
inductance for 40% ripple current is:
ΔI
V
fL
V
V
L
OUT OUT
IN
=
⎛
⎝
⎜
⎞
⎠
⎟
()( )
–1
A 14μH inductor will result in 40% ripple current. The peak
inductor current will be the maximum DC value plus one
half the ripple current, or 6A, for the 14μH value.
APPLICATIONS INFORMATION
The R
SENSE
resistor value can be calculated by using the
maximum current sense voltage specifi cation with some
accommodation for tolerances:
R
mV
A
SENSE
≤≈Ω
90
6
0 015.
Choosing 1% resistors; R1 = 20k and R2 = 280k yields
an output voltage of 12V.
The power dissipation on the top side MOSFET can be
easily estimated. Choosing a Siliconix Si4412DY results
in: R
DS(ON)
= 0.042Ω, C
RSS
= 100pF. At maximum input
voltage with T(estimated) = 50°C:
P
V
V
CC
MAIN
=
()
+°°
[]
12
30
5 1 0 005 50 25
0 042
2
(. )( – )
. ΩΩ
()
+
()()( )( )
=
1 7 30 5 100 250
664
2
.VA pF kHz
mW
A short-circuit to ground will result in a folded back
current of:
I
mV ns V
μH
A
SC
=
Ω
+
⎛
⎝
⎜
⎞
⎠
⎟
=
45
0 015
1
2
200 30
14
32
.
()
.
with a typical value of R
DS(ON)
and δ = (0.005/°C)(20) = 0.1.
The resulting power dissipated in the bottom MOSFET is:
P
VV
V
A
mW
SYNC
=
()()
Ω
()
=
30 12
30
32 11 0042
284
2
–
...
which is less than under full-load conditions.
Figure 9. Automotive Application Protection
V
IN
3727 F09
LTC3727A-1
TRANSIENT VOLTAGE
SUPPRESSOR
GENERAL INSTRUMENT
1.5KA24A
50A I
PK
RATING
12V
P1-P3P4-P6P7-P9P10-P12P13-P15P16-P18P19-P21P22-P24P25-P27P28-P30P31-P32

LTC3727AIG-1#PBF

Mfr. #:
Buy LTC3727AIG-1#PBF
Manufacturer:
Analog Devices / Linear Technology
Description:
Switching Voltage Regulators Dual, 2-Phase Synchronous Controller w/ up to 14V Output
Lifecycle:
New from this manufacturer.
Delivery:
DHL FedEx Ups TNT EMS
Payment:
T/T Paypal Visa MoneyGram Western Union

Products related to this Datasheet